HEWLETT-PACKARD JOURNAL

< * v

B u o y B u o y B u o y N o . 2 N o . 4 N o . 8

S A N F R A N C I S C O .

M E A S U R I N G T H E O C E A N ' S T E M P E R A T U R E : s e e p a g e 8 F R E Q U E N C Y D I V I D E R E X T E N D S A U T O M A T I C D I G I T A L F R E Q U E N C Y M E A S U R E M E N T S T O 1 2 . 4 G H z , p a g e 2 I M P R O V E D I N T E R M O D U L A T I O N D I S T O R T I O N R E J E C T I O N I N M I X E R S , p a g e 1 3

Cover See A lso

APRIL 1967 © Copr. 1949-1998 Hewlett-Packard Co.

Frequency Divider Extends Automatic Digital Frequency Measurements to 12.4 GHz T h i s m e a s u r e i n s t r u m e n t l e t s a n e l e c t r o n i c c o u n t e r m e a s u r e m i c r o w a v e f r equenc ies . The accu racy and s imp l i c i t y o f t he coun te r a re re ta i ned .

WHEN THE ELECTRONIC COUNTER was introduced in the early 1950's, the major advantages it offered

over existing techniques were simplicity and automatic operation. Tedious and complicated frequency measure ments were suddenly reduced to utter simplicity. Soon, devices to extend the upper frequency limit of the counter became available. Unfortunately, they were not auto matic and they were usually far from simple.

A new automatic frequency divider developed by -hp- successfully extends the frequency range of any electronic counter through X-band (12.4 GHz) while retaining the simple, automatic operation of the counter. Its basic prin ciple of operation is that of a prescaler which divides the

& Fig. 1. Automatic Frequency Divider, -hp- Model 5 260 A, divides input frequencies between 0.3 GHz and 12.4 GHz by 100 or 1000 so they can be measured by a counter (counter shown is -hp- Model 5245L 50 MHz Counter). Divider operation is com pletely automatic except for selection of division ratio.

input frequency by 100 or 1000. The reading on the coun ter requires only a decimal point shift for truly direct readout. The division ratio of 100 is used for frequencies below 1.2 GHz, and 1000 is used for frequencies above 1 GHz.

Fig. 1 , showing the front panel of the instrument, at tests to the simplicity of its operation. There is only one

knob on the front panel — the one for selecting the divi sion ratio. Once the division ratio has been chosen, the signal to be measured is applied at the input and the divided-down frequency appears at the output. A meter on the front panel indicates the approximate level (in dBm) of the input. The output, designed to drive a 50- ohm or higher load, features an automatic level control to maintain the output level at 0 dBm ± 2 dBm while the input varies over the specified range of — 7 dBm to +10 dBm.

Because of its automatic operation, the new frequency divider is ideally suited for use by relatively unskilled personnel in production test areas, or for use as a com ponent of a completely automatic, unattended system. Laboratories, too, will welcome the divider, because of the trend toward automated instruments which is now gaining momentum.1

Operation Although its front panel is the essence of simplicity,

the new frequency divider is actually a highly sophisti cated system. Its internal operation is quite unlike con ventional frequency dividers. It has the fixed division

1 For ano ther example o f the t rend towards au tomat ic ins t rumenta t ion , read about the new m ic rowave ne twork ana lyze r i n the Februa ry , 1967 i ssue o f t he 'Hew le t t -Packa rd Journal. '

P R I N T E D I N U . S . A .

© Copr. 1949-1998 Hewlett-Packard Co.

ratio of pulse-counting dividers, but demands a periodic input signal. It has frequency range capabilities similar to countdown dividers, but unlike these, it gives a constant division ratio. Unlike any other type of divider, it com bines a constant division ratio with microwave frequency capability.

A simplified block diagram of this new type of fre quency divider is shown in Fig. 2. The signal whose fre quency is to be divided is sampled by two wideband feedthrough samplers in series.- The pulses which open the sampler gates to cause periodic samples to be taken are derived from two voltage-tuned oscillators (VTO's) operating at approximately 30 MHz. Each VTO drives one of the samplers. The output of the first sampler is returned to the tuning input of the first VTO via the nec essary dc amplifiers and circuits to form an automatic phase-lock loop. When phase lock is established between the input frequency, fs, and the VTO frequency, fj, the two frequencies become harmonically related as shown in equation (1).

= nf , (1)

The frequency of the second VTO, f-, tracks the fre quency of the first VTO, f ,, with a slight offset. This offset results in a difference frequency at the output of the sec ond sampler. This frequency, fout, is n, the harmonic number, times the offset between the two VTO's. Thus,

f,,ut = n(f, - f,) (2 )

If the offset is chosen to be i J 100 or f,/1000 the out put frequency will be:

t o u t â € ” nf, 100 1000 ( 3 )

Substitution of equation (1) into equation (3) yields the desired result.

f o u t â € ” , 100

or t. 1000 (4)

The technique used to maintain the offset between f, and fo is shown in Fig. 3. The voltage which tunes the first VTO is fed forward to the second VTO. Since these VTO's are nominally identical, they will be driven to nearly the same frequency. This constitutes a coarse con trol on the tracking of the two oscillators. Fine control is provided by a phase-lock loop which compares the difference frequency between the two oscillators with the frequency of the first oscillator divided by 100 or 1000.

 ¡ T h e s e n e w s a m p l e r s , d e v e l o p e d b y h p a s s o c i a t e s , a r e a l s o u s e d i n a n e w - r i p - m i c r o w a v e n e t w o r k a n a l y z e r a n d  ¡ n a n e w - h p - s a m p l i n g o s c i l l o s c o p e . F o r a d e s c r i p t i o n o f t h e s a m p l e r s , s e e W . M . G r a v e , ' A D C t o 1 2 . 4 G H z F e e d t h r o u g h S a m p l e r f o r O s c i l l o s c o p e s a n d o t h e r R F S y s t e m s , ' ' H e w l e t t - P a c k a r d J o u r n a l , ' V o l . 1 8 , N o . 2 , O c t . , 1 9 6 6 . A n o t h e r a r t i c l e  ¡ n t h e s a m e i s s u e d e s c r i b e s t h e n e w s a m p l i n g o s c i l l o s c o p e : A . I . B e s t , D . L . H o w a r d , a n d J . M . U m p h r e y , ' A n U l t r a - w i d e b a n d O s c i l l o s c o p e B a s e d o n a n A d v a n c e d S a m p l i n g D e v i c e . ' F o r a d e s c r i p t i o n o f t h e n e t w o r k a n a l y z e r , s e e R . W . A n d e r s o n a n d 0 . t . D e n n i s o n , ' A n A d v a n c e d N e w N e t w o r k A n a l y z e r f o r S w e e p M e a s u r i n g A m p l i t u d e a n d P h a s e f r o m 0 . 1 t o 1 2 . 4 G H z , ' ' H e w l e t t - P a c k a r d J o u r n a l , ' V o l . 1 8 , N o . 6 , F e b . , 1 9 6 7 .

I n p u t f x

A U T O M A T I C P H A S E C O N T R O

CIRCUITS m = 1 0 0 o r 1 0 0 0

Fig. 2. Simplified block diagram of automatic frequency

divider. Control circuits maintain indicated relationships

between f,, f,, and f,.

F r o m M a i n Au toma t i c

Phase Cont ro l L o o p T o S a m p l e r T o S a m p l e r

Coarse F requency Con t ro l

Fig. 3. Block diagram of tracking control used to keep fre

quency offset (f, — f,) equal to f,/IOO or f,/1000.

© Copr. 1949-1998 Hewlett-Packard Co.

- 5 . 0 - 4 . 0 - 3 . 0 - 2 . 0 - 1 . 0 1 . 0 2 . 0

-1.0

- 3 . 0

- 4 . 0

-5.0

P, ' =0.72- i 2 .9

Fig. 4. (a). Magnified portion of root locus of main phase-

lock loop. Note poles in right half plane which migrate to

¡eft half plane as loop gain increases, (b) Vnmagnified root

locus of main loop. Right-half-plane poles shown in (a) appear to start from origin on this plot. As loop gain in

creases by 30 (IB, these poles move along line correspond

ing to damping ratios of 0.5 to 0.7. Thus loop is nearly

critically damped over wide dynamic range.

A correction voltage from the phase detector is ap plied to the second VTO, f:, through the necessary cir cuits to complete the phase-lock loop.

Design Features One of the major design features of the automatic fre

quency divider is the main phase lock loop, which enables the first VTO, f,, to lock onto the input signal without any intermediate frequency in the loop. Until now, micro wave phase lock loops have always used an intermediate frequency to establish the phase lock. This results in an offset between the microwave frequency and the har monics of the VTO so that an exact harmonic relation ship does not exist between fx and fj. The availability of the 12.4 GHz sampler has opened the door to zero-offset phase locking. Phase-related dc signals at a sufficiently high level to allow reliable phase locking are readily available from this sampler.

While the elimination of the intermediate frequency in the phase lock loop simplifies the system generally, one aspect becomes more complicated. In an IF system, input level variations are eliminated by the limiting action in the IF amplifier. With no IF, the amplitude information at the output of the phase detector (sampler) becomes indistinguishable from the phase information, and loop gain can be adversely affected. To overcome this diffi culty, some means must be used for independently meas uring the input signal amplitude and adjusting loop gain accordingly.

In the automatic frequency divider the input signal level is measured by detecting the level of the difference frequency out of the second sampler. The detected volt age is used to control the gain of the main phase lock loop. This same voltage is applied to the front panel meter to indicate the input signal level.

A disadvantage of previous microwave phase lock loops, which use harmonic mixers rather than samplers, has been the tendency of the harmonic mixer to generate harmonics of the input signal as well as harmonics of the local oscillator. The resulting phase locking between har monics of the local oscillator and harmonics of the input signal caused a large amount of uncertainty in frequency measurements. The use of a wideband sampler insead of a harmonic mixer has eliminated this problem because the sampler does not generate harmonics of the input signal.

Two major requirements placed upon the main phase lock loop of the frequency divider by the requirement for automatic operation of the instrument are (1) an auto matic search capability built into the loop to search out an input signal and lock onto it, and (2) a sufficiently wide gain margin for loop stability over a range of har monic numbers (n) from 1 0 to 400 (because loop gain is directly proportional to harmonic number).

© Copr. 1949-1998 Hewlett-Packard Co.

Frequency Divider + Integrated-Circuit Counter = 12.4 GHz Digital Frequency Meter

T h a n k s t o t h e s p a c e - s a v i n g h a b i t s o f i n t e g r a t e d c i r c u i t s , - h p - e n g i n e e r s h a v e m a n a g e d t o p u t a n a u t o m a t i c f r e q u e n c y d i v i d e r i d e n t i c a l t o t h e o n e d e s c r i b e d i n t h e a c c o m p a n y i n g a r t i c l e a n d a n e i g h t - d i g i t 1 C c o u n t e r i n t o o n e c o m p a c t p a c k a g e . T h e c o m b i n a t i o n m e a s u r e s a n y f r e q u e n c y b e t w e e n 0 . 3 G H z a n d 1 2 . 4 G H z , o r b e t w e e n 1 0 H z a n d 1 2 . 5 M H z , j u s t l i k e a n o r d i n a r y c o u n t e r . A l l t h e u s e r h a s t o d o i s c o n n e c t t h e u n k n o w n s i g n a l t o t h e p r o p e r i n p u t â € ” t h e r e ' s one for 0.3 to 12.4 GHz and one for 10 Hz to 12.5 MHz — and s e l e c t t h e p r o p e r f r e q u e n c y r a n g e . E v e r y t h i n g e l s e i s a u t o m a t i c . T h e e i g h t - d i g i t r e a d o u t i n c l u d e s t h e d e c i m a l p o i n t and measu remen t un i t s . The re i s a l so a p r i n t e r ou tpu t (BCD i s s tanda rd ) w i t h dec ima l po in t and un i t s .

F o r m e a s u r i n g t h e h i g h f r e q u e n c i e s , t h e d i v i d e r l o c k s o n t o t h e i n p u t s i g n a l a n d s u p p l i e s t h e 1 C c o u n t e r w i t h a s i g n a l w h i c h h a s 1 / 1 0 0 o r 1 / 1 0 0 0 o f t h e i n p u t f r e q u e n c y . B o t h c o u n t e r r e a d o u t a n d p r i n t e r p r i n t o u t a r e p r e v e n t e d u n t i l t h e

- h p - 5 2 4 0 A 1 2 . 4 G H z D i g i t a l F r e q u e n c y M e t e r

f r e q u e n c y d i v i d e r h a s l o c k e d o n t o t h e i n p u t s i g n a l . W h e n low f requenc ies a re be ing measured , the d i v ide r i s bypassed a n d t h e 1 C c o u n t e r m e a s u r e s t h e u n k n o w n f r e q u e n c y d i r e c t l y . ( I n c i d e n t a l l y , m o s t o f t h e I C ' s f o r t h e c o u n t e r w e r e p roduced in -hp - ' s own 1C fac i l i t i es . )

S p e c i f i c a t i o n s f o r t h e 1 2 . 4 G H z D i g i t a l F r e q u e n c y M e t e r are on p. 7.

The automatic search feature is achieved by inserting a low-frequency oscillator into the loop and adjusting the loop compensation so that the loop itself oscillates when loop gain is very low, and the low-frequency oscillator oscillates whenever the loop is open (unlocked). The oscillations sweep the first VTO through its frequency range until an input signal is found and phase lock is established.

A magnified portion of the root locus of the main phase lock loop is reproduced in Fig. 4(a). This shows the existence of poles in the right half plane which mi grate to the left half plane as loop gain increases. The right-half-plane poles help to eliminate locking to spuri ous lower-level signals present along with the input signal. Since the loop is unstable for the low gain ensuing from a poor lock point it will automatically search out the best lock.

The locus of the poles as loop gain is increased still further as shown in Fig. 4(b). Here it can be seen that the poles move along a line maintaining a damping ratio of C = 0.5 to ç = 0.7 as loop gain increases by 30 dB. Thus, with these poles dominating the system response, the loop is nearly critically damped over a very wide dynamic range.

Performance Overall performance of the automatic frequency di

vider can be judged by two factors. First, how well does it lock on to input signals with amplitude or frequency modulation? Second, how well can it produce a signal at

Max imum Modu la t i on Ava i l ab l e f r om S igna l Genera tor Used in Tes t

£ looo

o LlJ CE

100

0 . 1 1 1 0

MODULATING FREQUENCY fm (kHz)

100

Fig. 5. Maximum frequency modulation on input signal

that will permit frequency divider to lock on. These are

typical measured characteristics at 0 dBm input. Division

ratio was 1000 except where indicated.

© Copr. 1949-1998 Hewlett-Packard Co.

100

80 z O 5 6 0

§ 40 5

^ 2 0

M a x i m u m M o d u l a t i o n A v a i l a b l e 0 .3 GHz + loo

0 . 0 1 0 . 1 1 . 0 1 0 1 0 0 1 0 0 0

M O D U L A T I N G F R E Q U E N C Y f m ( k H z )

Fig. 6. Maximum amplitude modulation on input signal

that will permit frequency divider to lock on. These are

typical measured characteristics at 0 dBm input.

Fig. 7. For AM signals, frequency divider output tends to

be a lower-frequency replica of the input . Traces show

div ider output for 1 GHz input s ignal , 20% ampl i tude

modulated by 100 kHz signal. Top: divider output for divi

sion ratio m — ¡00. Bottom: divider output for m — 1000.

its output which is suitable to drive an electronic counter, even though the input signal may be amplitude or fre quency modulated?

Fig. 5 is a plot of the maximum FM deviation and FM rate which can be tolerated by the main phase lock loop before it will fail to achieve phase lock. Each of the curves in Fig. 5 is an experimental curve for a single sinusoidal input signal, frequency modulated by a single sinusoidal signal of frequency fm. (The rms frequency deviation Af is proportional to the amplitude of the mod ulating signal.) These curves show, for example, that for an input frequency of 12 GHz and a modulating fre quency of 1 kHz, the main loop will not achieve phase lock for rms frequency deviations greater than 480 kHz.

However, when the modulating frequency is reduced to 1 00 Hz, the maximum allowable rms frequency deviation increases to 2.6 MHz.

If an input signal is frequency modulated by a non- sinusoidal signal, the frequency fm and amplitude Af of each component of the modulating signal must be taken into account. The effects of all components will be addi tive, so the performance of the frequency divider for each component will be degraded somewhat from that shown in Fig. 5. A detailed mathematical analysis of this prob lem is still in progress, so it is not yet possible to say precisely how the components of a nonsinusoidal signal interact in the phase-lock loop.

Robert L. Allen

B o b A l l e n r e c e i v e d h i s B S d e g r e e i n e l e c t r i c a l e n g i n e e r i n g f r o m U t a h S t a t e U n i v e r s i t y i n 1 9 6 0 . H e w o r k e d t h e f o l l o w i n g s u m

m e r f o r t h e - h p - F r e q u e n c y a n d T i m e D i v i s i o n , a n d t h e n r e t u r n e d t o U t a h S t a t e , g r a d u a t i n g i n 1 9 6 1 w i t h a n M S d e g r e e i n e l e c t r i c a l e n g i n e e r i n g . I n S e p t e m b e r , 1 9 6 1 , h e j o i ned t he F requency and T ime D i v i s i on f u l l - t ime . He con t r i b u t e d t o t h e d e s i g n o f t h e 5 2 7 5 A a n d 5 2 4 3 L C o u n t e r s a n d t h e 1 0 7 A Q u a r t z O s c i l l a t o r , a n d w a s p r o j e c t l e a d e r f o r d e v e l o p m e n t o f t h e 5 2 6 0 A F r e q u e n c y D i v i d e r a n d t h e 5 2 4 0 A D i g i t a l F r e q u e n c y M e t e r . H e h a s a p a t e n t p e n d i n g a n d h a s p u b l i s h e d a p a p e r o n t h e m e t h o d o f f r e q u e n c y d i v i s i o n u s e d i n t h e 5 2 6 0 A a n d 5 2 4 0 A .

B o b i s a m e m b e r o f I E E E , S i g m a X i , P h i K a p p a P h i , a n d S i g m a T a u . B y t h e t i m e t h i s a r t i c l e a p p e a r s i n p r i n t , h e w i l l h a v e a s s u m e d n e w r e s p o n s i b i l i t i e s a s e n g i n e e r i n g m a n a g e r o f t h e - h p - D e l c o n D i v i s i o n i n M o u n t a i n V iew , Ca l i f o rn ia .

© Copr. 1949-1998 Hewlett-Packard Co.

Incidental FM or phase noise on the input signal and phase noise generated in the divider also affect perform ance, but since the curves of Fig. 5 are experimental, they include the effects of these noise signals.

Regardless of what kind of modulating signal is pres ent, once the frequency divider locks onto an FM signal, a counter on the divider output will read the carrier fre quency correctly. The modulation will be averaged out. Therefore, for FM signals, the answer to the second ques tion stated above follows from the answer to the first, that is, if the divider can lock onto a signal, the divider output is suitable to drive a counter.

Amplitude modulation on an input signal affects the locking performance of the frequency divider as shown in Fig. 6. These curves represent experimental data for sin usoidal input signals, amplitude modulated by sinusoidal signals of frequency fm. The ordinate is percentage am plitude modulation. Thus, the divider will lock onto an input signal at 12.4 GHz which is amplitude modulated up to 47% by a 200 kHz signal. However, if the modu lating frequency fm is 100 kHz or less, the maximum al lowable AM is 30%. The increased immunity to AM at high modulation rates is caused by the rolloff in the fre quency response of the main phase-lock loop. This same effect can be seen in the FM curves of Fig. 5.

Since the divider output is derived from a sampler, the output signal tends to be a lower frequency replica of the input signal. When the input signal is amplitude modu lated, so is the output signal. To demonstrate this, a di vider was supplied with a 1 GHz carrier, 20% amplitude modulated by a 100 kHz signal. Fig. 7 shows the cor responding output for division ratios of 100 and 1000, respectively.

If the AM causes the troughs in the envelope of the output signal to dip below the sensitivity limits of a coun ter connected to the output, miscounting will occur. For a counter with 100 mV sensitivity, this occurs with ap proximately 50% AM.

Acknowledgments My associates in the electrical design of the 5260A

Frequency Divider were Achule M. Bigliardi, Glenn B. DeBella. Rolf B. Hofstad, Ronald D. Lowe, Richard F. Schneider, and Ronald K. Tuttle. The electronic counter and additional circuits for the 5240A Digital Frequency Meter were designed by Ronald D. Lowe. Richard W. S. Goo served as a technician on both projects and his as sistance is much appreciated. Mechanical design for both instruments was the work of Glen E. Elsea, a 10-year veteran at -hp-. Richard F. Schneider, who designed the zero-offset phase-lock loop, deserves special credit for his analysis and evaluation of the divider as a system and of its performance with modulated input signals. •

S P E C I F I C A T I O N S

- h p - M o d e l 5 2 6 0 A

F requency D iv ide r

RANGE: 0 .3 to 12 .4 GHz A C C U R A C Y : R e t a i n s a c c u r a c y o f e l e c t r o n i c c o u n t e r . INPUT SENSIT IV ITY : 100 mV rms (†”7 dBm) . I N P U T I M P E D A N C E : 5 0 o h m s n o m i n a l .

Freq. 0 .3-8 GHz 8 -10 GHz

10-12.4 GHz

I N P U T V S W R T y p i c a l

1.2:1 1,4:1 1.8:1 ,

: l e v e l 1 0 d B m M A X I M U M I N P U T : 1 0 d B m L E V E L I N D I C A T O R : F r o n t p a n e l m e l e r i n d i c a t e s a p p r o x i m a l

t o - f 1 0 d B m . D I V I S I O N R A T I O : F r o n t p a n e l s w i t c h s e l e c t s â € ” 1 0 0 ( f o r u s e u p t o 1 . 2 G H z ) o r ^ 1 0 0 0

( f r o m 1 t o 1 2 . 4 G H z ) o p e r a t i o n . I N P U T C O N N E C T O R : P r e c i s i o n T y p e N f e m a l e . O P E R A T I O N : C o m p l e t e l y a u t o m a t i c o n c e t h e D I V I S I O N R A T I O s w i t c h i s p o s i t i o n e r ) O U T P U T F R E Q U E N C Y : 1 / 1 0 0 o r 1 / 1 0 0 0 o ( i n p u t ( 1 t o 1 2 . 4 M H z ) . O U T P U T I M P E D A N C E : D e s i g n e d f o r 5 0 o h m l o a d . O U T P U T L E V E L : 0 d B m . n o m i n a l . R E G I S T R A T I O N : I n p u t f r e q u e n c i e s f r o m 0 . 3 t o 1 2 4 G H z a r e m e a s u r e d b y m e a s u r i n g

t h e 5 2 6 0 A o u t p u t w i t h a n e i e c t r o n i c c o u n t e r s u c h a s t h e ~ h p - 5 2 4 5 L , 5 2 4 6 L o r 5 2 4 4 L , a n d s u i t a b l y p o s i t i o n i n g t h e d e c i m a l p o i n t . R e a d o u t i s d i r e c t w i t h n o o f f s e t , a m b i g u i t y o r a r i t h m e t i c p r o c e s s i n g . S e e a l s o O p t i o n 0 2 . b e l o w .

P R I C E : M o d e l 5 2 6 0 A A u t o m a t i c F r e q u e n c y D i v i d e r , S 3 4 5 0 . OPTIONS:

0 1 . A m p h e n o l A P C - 7 I n p u t C o n n e c t o r , a d d $ 2 5 . 0 0 . 0 2 . P r o v i d e s 5 2 6 0 A w i t h c i r c u i t r y s u c h t h a t , w h e n u s e d w i t h t h e - f t p - M o d e l M O 7 - 5 2 4 5 L E l e c t r o n i c C o u n t e r , t h e d e c i m a l p o i n t w i l l b e a u t o m a t i c a l l y p o s i t i o n e d f o r r e a d o u t i n G H z , a n d t h e s y m b o l ' G H z * w i l l a p p e a r i n t h e c o u n t e r ' s r e a d o u t . R e a d o u t s u p p r e s s e d w h e n d i v i d e r u n l o c k e d . A d d 5 1 7 5 . 0 0 .

t e m p e r a t u r e : < Â ± 2 + 6 5 * C ) . A s f u n c t i o n

M o d e l 5 2 4 0 A 1 2 . 4 G H z D i g i t a l F r e q u e n c y M e t e r

F R E Q U E N C Y D I V I D E R P O R T I O N ( S p e c i f i c a t i o n s s a m e a s 5 2 6 0 A )

L O W - F R E Q U E N C Y C O U N T E R P O R T I O N

F R E Q U E N C Y M E A S U R E M E N T S : RANGE: 10 Hz t o 12 .5 MHz . G A T E T I M E : 0 . 1 . 1 . 0 s : 1 0 s o n s p e c i a l o r d e r . S E L F C H E C K : C o u n t s 1 M H z f o r g a t e t i m e c h o s e n .

S I G N A L I N P U T ; S E N S I T I V I T Y : 1 0 0 m V r m s . M A X . I N P U T : 2 V r m s . I M P E D A N C E : 1 M i ? s h u n t e d b y 1 0 p F .

T I M E B A S E : •FREQUENCY: 1 MHz.

S T A B I L I T Y : A g i n g r a t e : < 2 p a r t s i n 1 0 ' / m o n t h . A s f u n c t i o n o f p a r t s i n 1 0 * ( + 1 0 Â ° C t o + 5 0 Â ° C ) , < Â ± 2 0 p a r t s m 1 0 * ( 0 " C t o o f l i ne vo l tage ( 11 10%) : 1 par t in 107 . EXTERNAL INPUT: 1 V rms in to 1 k1 . . ! . OUTPUT: 1 MHr , 2 V squa re wave i n to 6 k ' . . ' .

R E M O T E R E S E T : B y g r o u n d i n g c e n t e r o f B N C b a c k p a n e l .

GENERAL

A C C U R A C Y : 1 c o u n t  ± ; t i m e b a s e a c c u r a c y . R E A D S I N : M H z o r G H z w i t h p o s i t i o n a l d e c i m a l p o i n t . P R I N T E R O U T P U T : C o m p a t i b l e w i t h - r i p - M o d e l s 5 6 2 A a n d 5 0 5 0 A D i g i t a l R e c o r d e r s

w i t h 1 - 2 - 4 - 9 B C D ' V s t a t e n e g a t i v e ( a n d O p t i o n 1 4 f o r t h e 5 6 2 A ) . P r i n t e r s r e c o r d d e c i m a l p o i n t a n d m e a s u r e m e n t u n i t s ( 5 6 2 A o n l y ) .

C H A S S I S C O N N E C T O R : A m p h e n o l o r C i n c h t y p e 5 7 - 4 0 5 0 0 - 3 7 5 . - f t p - p a r t n o . A S I Ã “ O S " . 5 0 p i n . f e m a l e . M a t i n g c o n n e c t o r A m p h e n o l o r C i n c h t y p e 5 7 - 3 0 5 0 0 - 3 7 5 . - h p - pa r t no . 1251-0086 , 50 p in . ma le .

•0' STATE LEVEL: -6 V ' 1 ' S T A T E L E V E L : 0 V IMPEDANCE: 20 k ' . . ' . each l i ne . B C D R E F E R E N C E L E V E L S :

GROUND: i 5 V . 1 k ' . .1 source . P R I N T C O M M A N D : 0 V t o 1 0 V s t e p , d c c o u p l e d . H O L D - O F F R E Q U I R E M E N T S : - 1 5 V m a x . . i 2 . 5 V m i n .

P R I C E : M o d e l 5 2 4 0 A . $ 4 7 5 0 . M A N U F A C T U R I N G D I V I S I O N : ~ h p - F r e q u e n c y a n d T i m e D i v i s i o n

1 5 0 1 P a g e M i l l R o a d Pa lo A l t o , Ca l i f o rn i a 94304

— Robert L. Allen

© Copr. 1949-1998 Hewlett-Packard Co.

Precision Measurement of Ocean Temperatures A s o c e a n r e s e a r c h b e c o m e s m o r e s o p h i s t i c a t e d , g r e a t e r p r e c i s i o n i n t e m p e r a t u r e m e a s u r e m e n t i s n e e d e d

BENJAMIN FRANKLIN was able to locate and chart the Gulf Stream by hauling buckets of water on deck

and measuring its temperature. Today, many northbound ships locate themselves in the Gulf Stream by towing thermometers, then take advantage of the current. This is only one of many uses for ocean temperature data.

Life in the sea, ocean currents, tidal changes, acoustic transmission, radiation absorption and heat loss are re flected by even minute ocean temperature variations. Oceanographic instruments for measuring other param eters are generally calibrated on the basis of temperature data. All of these temperature measurements require varying degrees of precision — some of them to within a millidegree C or F.

Temperature patterns are three dimensional, and both horizontal and vertical distributions influence the move ment of fish. Changes of only 3 or 4 degrees C are a large change to these cold-blooded animals and have a marked effect on their metabolism and activity. In many areas, fishermen rely on temperature information to increase the size of their catch.

Temperature differences at all depths generate ocean currents, and these differences are used by oceanogra- phers in the study of currents. The same temperature

variations influence the propagation of sound through the water, and hence influence the effectiveness of sound ing, tracking and detection devices.

Because of the variety of applications of ocean tem perature data, many different types of measurements are required. The most common is a measurement of the thermocline, the temperature gradient from the surface layer to the essentially isothermal deep water. Depths may range to hundreds of meters, and temperature differences may be as high as 20 degrees C. This measurement per mits a prediction of sound velocity with depth and how sonar beams will deflect when directed into the water, Fig. 1.

Another important temperature gradient measure ment extends over distances of one or two meters above and below the water surface. Temperature gradients of 10 millidegrees per meter or less indicate the vertical flux of heat from water to air. Most solar radiation is absorbed by the oceans of the world and converted to heat in the water. Some of this heat is transferred to the atmosphere by winds and is a primary influence on weather. These measurements are vital to the study of climate and will be used in long-range weather fore casting.

(a)

5 0 Â ° 7 0 Â °

Fig. 1. Bending of ti trtuiMHilteil atinar beam la} is caused

by a combination of ti xnulual temperature elimine mid a

cliange in density of I/if water. At certain limes <>j day.

and ajter t i s torm, m'txini: occurs at the surface which

results in a constant t em fie ram r c zone. I 'inter this condi

tion, a sonar beam launched at an anule may return to the

50°

surface fb). Since the beam is influenced only by a change

in density and not temperature, the speed of the lower part

of the wavejront is higher than the upper part causing the

beam to continue to bend. An object, such as a submarine

in this area may escape detection. Precise temperature

measurements can detect these temperature gradients.

© Copr. 1949-1998 Hewlett-Packard Co.

Fig. 2. Temperature sensor

assembly is being mounted on an

instrument tripod. It can be

submerged to depths over

20,000 feet and can make

temperature measurements l<> a

resolution of 1/10 milliilcnree.

The measurements of salinity, depth and velocity are made with transducers which, in common with most sensors, have significant temperature coefficients which affect both their zero setting and their sensitivity. When very precise measurements are to be made, these coeffi cients must be determined and the necessary corrections made on the basis of the indicated 'in-situ' temperature.

An excellent example of the use of temperature meas urements is in the study of tides being conducted both by the Institute of Geophysics and Planetary Physics at the Scripps Institute of Oceanography in La Jolla, Cali fornia, and the Institute of Geophysics at the University of Hawaii. Until recently man was able to measure tidal changes only at the edges of an ocean's basin and on islands where he could determine tidal height with refer ence to the solid earth. This limited the amount of data available for a complete study of the tidal mechanism and made it difficult to locate key nodal points of the tidal oscillations.

Scripps Institute has been measuring the tide at se

lected points in the Pacific Ocean by dropping complete instrument packages to the bottom to measure and record the changes in water pressure and temperature, Fig. 2, for periods as long as several months. The indicated pres sure is a function of water height at the observation point but is also influenced by changes in water density, which can be computed with a knowledge of the temperature. The pressure sensor output which is also a function of temperature is corrected as well to determine the true tidal variations. Since the height is required to an accu racy of better than one foot and since most off-shore locations in the oceans are at depths of 10.000 to 15,000 feet, a precision of at least 0.01 % is required.

Instrument Design Problems Reliability and accuracy must be available at reason

able cost to satisfy oceanographers. Seawater corrosion, temperature variations, high pressures and high humidity contribute to severe environmental conditions that create problems common to all oceanographic instruments. For

© Copr. 1949-1998 Hewlett-Packard Co.

J Fig. 3. A temperature sensing probe, the —hp— Model 2832 A , top, houses the quartz crystal and all electronics in a pressure ca.se designed .specifically for measurement of

ocean temperatures. At the bottom, the —hp— Model 2833A sensor assembly in a .stainless steel case will with stand pressures to 10,000 psi and may be operated at

distances up to one mile from the temperature readout.

Fig. 4. A complete quartz thermometer .system, the —hp— Model 2801 A with two temperature sensing probes. With this system, the temperature of either probe or the differ ence between the temperatures of the two probes can be shown on the readout.

example, it is necessary to maintain water-tight integrity under pressures as high as 1 5,000 to 20,000 psi. Ther mometers, in addition, have the problem of self-generated heat which causes significant offsets in the order of milli- degrees when located where currents are near zero. Re sponse time, which determines how rapidly a thermo- metric probe can be lowered and still detect abrupt changes, is also affected by self-generated heat.

Effects of self-generated heat may be reduced by de signing the probe such that the sensor is separated from the heat generating components. Choice of appropriate materials and careful mechanical design have done much to alleviate other problems.

Data Transmission The oceans and their characteristics will be completely

revealed only when data gathering systems provide syn optic data so that man can see the whole 300 X 10'1 cubic miles simultaneously. There is a problem then, in gather ing data from many sources.

Multiple strings of electrical sensors can be employed on ships, from moored buoys, and from bottom-planted systems. While the data being sensed is analog, it is usu ally converted to digital form for use with computing systems and digital displays. An intermediate step com monly employed to permit the telemetering of the data from sensor to system is to convert to frequency modu lation immediately after the sensor to reduce inaccura cies possible in analog transmission. Then the data from individual sensors must be transmitted fairly rapidly to a central location to be reduced, correlated and plotted. However, operating systems of this type are somewhat in the future.

Quartz Thermometer Prior to the introduction of the quartz sensor either

resistance elements or thermistors were used. These may be biased with a constant current to produce a dc voltage to operate a voltage-controlled oscillator, or used in phase shift networks to influence the frequency of an RC oscil lator. In either case the stability of the frequency conver sion is influenced by other circuit elements in addition to the sensor itself.

The quartz temperature sensor takes advantage of the characteristic of a quartz crystal to shift its frequency of oscillation with temperature.' •- The sensor is a self-con tained temperature-sensitive oscillator that requires only microwatts of power to maintain oscillation. The LC cut sensor has a precisely linear frequency-to-temperature relationship and has a sensitivity of approximately 1000 Hz per degree Centigrade.

The simplest model, Fig. 3, is the -hp- 2833A Tem-

1 . A l b e r t B e n j a m i n s o n ' T h e L i n e a r Q u a r t z T h e r m o m e t e r , ' ' H e w l e t t - P a c k a r d J o u r n a l ' V o l . 1 6 , N o . 7 , M a r . , 1 9 6 5 , p . 2 . 2 . D . L i n e a r ' A C . A . A d a m s , P . S c h m i d t , H e w l e t t - P a c k a r d C o . , ' A L i n e a r Q u a r t z C r y s t a l T e m p e r a t u r e S e n s i n g E l e m e n t , ' p r e s e n t e d a t t h e 1 9 t h A n n u a l I S A C o n f e r e n c e , O c t . 1 2 - 1 5 , 1 9 6 4 , N e w Y o r k . P r e p r i n t N o . 1 1 , 2 - 3 - 6 4 .

10

© Copr. 1949-1998 Hewlett-Packard Co.

Surface Temperature Profile of the Ocean

Surface Surface

Mixed Zone Negat ive Grad ien t

Layer Depth ' Layer Depth t

TEMPERATURE (a)

TEMPERATURE (b)

Posi t ive Grad ien t

Layer Depth

Sound Channel

TEMPERATURE ( C )

T h e o c e a n s a r e c h a r a c t e r i z e d b y t h r e e d i s t i n c t l a y e r s : ( a ) a n u p p e r l a y e r n o r m a l l y i s o t h e r m a l d u e t o m i x i n g b y w i n d s , a m i d d l e ( t h e r m o c l i n e ) r e g i o n o f s t r o n g n e g a t i v e g r a d i e n t s , a n d a l o w e r r e g i o n o f s l o w c h a n g e e x t e n d i n g t o t h e b o t t o m ; ( b ) i l l u s t r a t e s t h e ' a f t e r n o o n e f f e c t ' w h e n s u m m e r h e a t a n d l i g h t w i n d s c a n c a u s e a n e g a t i v e g r a d i e n t i n t h e u p p e r l a y e r . A p o s i t i v e g r a d i e n t ( c ) c a n a l s o f o r m w h e n c o o l ,

n o n - s a l i n e s h e l f w a t e r f l o w s o v e r t h e w a r m , s a l i n e G u l f S t ream.

I n s o m e a r e a s , t h e t e m p e r a t u r e b e l o w t h e t h e r m o c l i n e dec reases w i th dep th , t hen inc reases aga in , f o rm ing a sound c h a n n e l a t t h e m i n i m u m t e m p e r a t u r e d e p t h i n w h i c h l o n g s o n a r r a n g e s a r e p o s s i b l e , m a k i n g t h i s c h a r a c t e r i s t i c v e r y impor tan t .

perature Sensor Assembly which contains within a small stainless steel pressure case, a sensor, a solid state oscil lator and an amplifier. A coaxial cable feeds dc to the unit and returns a frequency-modulated RF signal to the data acquisition instrument or system. These sensors are used singly and in arrays to provide continuous data at selected sites. Since the output frequency is determined almost entirely by the sensor itself, cable length, and tem perature, or small changes in resistance or supply voltage have negligible effect on the signal. It can be transmitted 5000 feet or more without loss of accuracy. The RF sig nal is normally compared with a reference frequency and the difference frequency normalized and displayed in degrees C.

When very high stability is desired, the reference crys tal and its companion oscillator may be placed in the same case or in duplicate cases. The AT cut reference crystal is similar to the LC cut sensor except that its temperature coefficient is only a few Hz per degree C compared with the 1000 Hz per degree C of the sensor. By mixing the signals, one benefits both from an overall calibration of the system and the advantages of transmit-

over S h o v i n g o f f a t 4 : 0 0 o n a v e r y w i n d y A p r i l m o r n i n g , A l B e n j a m i n s o n a n d a c r e w o f t h r e e s a i l e d A l ' s 2 9 - f o o t c u t t e r ' W e n d y A n n ' 2 6 m i l e s t h r o u g h t h e G o l d e n G a t e t o t h e F a r a l l à ³ n I s l a n d s . A c o n t i n u o u s p l o t o f s u r f a c e t e m p e r a t u r e w a s m a d e u s i n g a n - h p - M o d e l 2 8 5 0 B Q u a r t z T h e r m o m e t e r m o u n t e d i n a t h r o u g h - h u l l f i t t i n g . T h e 6 0 - m i l e round t r i p l as ted abou t 12 hou rs .

ting a smaller difference frequency to a data system or low frequency counter.

Many variations can be made of these basic combina tions. Two sensor signals, Fig. 4, can be mixed to pro vide differential temperature readings without resorting to a reference comparison, or both absolute and differential readings can be obtained either simultaneously or sequen tially. For example, in making temperature gradient measurements, one thermometer may be kept at the sur face while the other is submerged. The difference may

Fig. 5. Where a probe is used for precision measurements

in s t i l l water , heat diss ipated by the electronics in the

package may heat the surrounding water, thus influencing

the probe temperature. The modification shown here places

the quartz crystal at some distance from the remainder of

the probe and minimizes this effect.

11

© Copr. 1949-1998 Hewlett-Packard Co.

Fig. the University torpedo used by the Applied Physics Laboratory of the University

of Washington for oceanographic research. A precision thermometer sensor mounted in the nose

enables the vehicle to follow constant temperature lines through the water.

be read directly. Switching and scanning systems are made to read the sensors in sequence either manually or automatically. Since small changes in resistance or ther mal contact voltages do not influence the accuracy, con siderable freedom in switching design can be realized.

Because even the small power dissipation of a sensor assembly may create a disturbing temperature gradient around the case (as it might for extremely high accuracy measurements at the deep ocean bottom) it is practical to extend the sensor in front of the case as shown in Fig. 5, and use the larger case to provide more surface area for heat dissipation.

On a typical oceanographic vessel which may cruise to many stations and require a variety of temperature data, different types of assemblies can be used simultaneously depending upon where they are to be installed. The -hp-

2850C probe can be installed in the condenser water in take and in the pilot log chamber, or other thru-hull fitting, usually in the deepest part of the bow. These per mit continuous display, both graphical and digital, of these temperatures.

Other probes can be installed in shielded aspirators to monitor air temperatures at various levels above the sea's surface. A probe installed in a fixed station package per mits vertical profiling of temperature in conjunction with salinity, pressure and other parameters of interest. A probe is frequently installed in a 'fish' and towed behind a vessel for more accurate information on sea surface temperatures. All of these probes can be connected to a central data acquisition system and scanned manually or automatically.

An interesting application of temperature sensing is made at the Applied Physics Laboratory, University of Washington, to a telechiric system (remote-controlled robot) which consists of an automatic torpedo-like 'fish,'

Fig. 6, that obeys commands sent by sonar from a surface vessel to start, stop, turn and change speed. It can be independent in vertical movement, however. The elevator mechanism is controlled either by a pressure or tempera ture sensor. Under these controls the 'fish' then traces the vertical pattern of an isobar (constant pressure line) or isotherm (constant temperature line), and its movements are followed by the fathometer on board the mother ship. In this way the isobar or isotherm can be completely charted, providing further information on currents, dens ity changes and internal waves which are so vital to our growing knowledge of the sea and its parameters. •

— A Ibcrt Benjaminson

Albert Benjaminson

A l B e n j a m i n s o n i s a g r a d u a t e o f t h e U n i v e r s i t y o f A d e l a i d e , S o u t h A u s t r a l i a , w i t h t h e d e g r e e o f B E E . H e h a d p r e v i ous ly a t tended the Po ly techn ic I n s t i t u t e o f B r o o k l y n . B e f o r e j o i n i n g t h e H e w l e t t - P a c k a r d D y m e c D i v i s i o n i n 1 9 5 9 h e h a d w o r k e d a s a d e s i g n e n g i n e e r o n c o m m e r c i a l r a d i o p r o d u c t s a n d w a s a l s o a s e c t i o n h e a d f o r m i l i t a r y a i r bo rne e l ec t ron i c sys tems .

S i n c e j o i n i n g - h p - , h e h a s b e e n E n g i n e e r i n g M a n a g e r o f R F S y s t e m D e v e l o p m e n t a n d w a s r e s p o n s i b l e f o r t h e d e v e l o p m e n t o f t h e p h a s e - l o c k e d o s c i l l a t o r s y n c h r o n i z e r , a m i c r o w a v e f r e q u e n c y c o n v e r t e r a n d a t u n a b l e V L F r e c e i v e r . I n 1 9 6 4 h e w a s m a d e e n g i n e e r i n g m a n a g e r f o r t r a n s d u c e r d e v e l o p m e n t a n d i s r e s p o n s i b l e f o r q u a r t z t h e r m o m e t e r d e v e l o p m e n t .

12

© Copr. 1949-1998 Hewlett-Packard Co.

Improved Intermodulation Rejection in Mixers I n t e r m o d u l a t i o n d i s t o r t i o n , a l w a y s a p r o b l e m i n m i x e r d e s i g n , c a n b e l a r g e l y p r e v e n t e d b y a c a r e f u l c h o i c e o f b i a s a n d power leve ls .

INTERMODULATION has always been one of the most per nicious forms of spurious response in superheterodyne receivers. It is a pleasure to be able to report that this type of interference can now be vastly reduced by proper design of the mixing element of the receiver and by a proper choice of the dc operating points of the mixer diodes.

By means of the recently discovered techniques de scribed in this article, intermodulation (IM) rejection has been improved from the 50 dB considered good in con ventional mixers to over 80 dB in a single-ended hot- carrier-diode mixer and to over 100 dB in a balanced hot- carrier-diode mixer.1 Moreover, this tremendous im provement in IM rejection has been obtained with little or no loss of sensitivity and little or no effect on rejection of other spurious responses. The new technique is not limited to hot carrier diodes; it can be applied to other mixing devices, and has been successfully applied in the design of a mixer using insulated-gate field-effect tran sistors.

Causes of Intermodulat ion Inputs to a conventional mixer are 1) a fixed-frequency

local-oscillator signal and 2) the receiver input signals, which may have frequency components anywhere in the passband of the receiver input. The mixer, a square-law device, raises the sum of the input signals and the local- oscillator signal to the second power, thereby generating cross-products, some of which have frequencies that are the differences between the input frequencies and the local-oscillator frequency. These difference-frequency components are the desired outputs; other cross-products are eliminated by a bandpass intermediate-frequency (IF) amplifier.

i J . H . Lepo f f and A . M. Cowley , ' Improved In te rmodu la t ion Re jec t ion in M ixe rs , ' IEEE T ransac t i ons on M ic rowave Theo ry and Techn iques , Vo l . MTT-14 , No . 12 , December , 1966.

Intermodulation results when harmonics of two or more incoming frequency components combine with har monics of the local oscillator frequency to produce re sponses which fall within the passband of the IF ampli fier. These harmonics are generated in any mixer which is not a perfect square-law device, or in other words, in any real mixer.

Spurious IM responses result when the sum of two input components and the local oscillator signal is raised to the fourth or higher power. In a diode mixer there are usually several significant higher power coefficients in the power series expansion of the diode current-vs-voltage function. This power series expansion about the dc op erating point of the diode is

j= oo

where

= o

i(v) = diode current, v = diode voltage,

v0 = diode bias voltage, or dc operating point,

V =

The coefficient k= is the one that gives the desired square- law mixing. Terms for which j is an odd number (i.e., k1;

100 r—

E 6 0 -

1 2

VOLTAGE v (V)

Fig. curve Theoretical hot-carrier-diode curve and measured curve for typical hpa 2300-series diode.

1 3

© Copr. 1949-1998 Hewlett-Packard Co.

-. 10

L O C A L O S C I L L A T O R P O W E R ( d B m )

Fig. 2. Intermodulation rejection of a single-ended hot-carrier-

diode mixer. Diode bias was adjusted for best ¡M rejection at

each local-oscillator level. 1M rejection can be maximized by

proper selection of diode bias and LO power, as explained in text.

110

1 4 1 8 2 2 2 6

L O C A L O S C I L L A T O R P O W E R ( d B m )

Fig. Diode IM rejection of a balanced hot-carrier-diode mixer. Diode

bias vol tages were adjusted for best IM reject ion at each LO

level . For LO power above 18 dBm, maximum desired-signal

power was used, and IM was below mixer noise level.

k3, k5, etc.) do not cause trouble because they do not pro duce cross-product frequencies in the IF passband. Inter- modulation is caused by non-zero values of k4, k, k., etc., in the current-vs-voltage characteristic of the diodes.

Hot-Carr ier -D iode Mixers The current-vs-voltage characteristic of a typical hot-

carrier diode is shown in Fig. 1 . The top curve in Fig. 1 is calculated from the expression

where is

Rs

Ue'(TUlV— 1].

saturation current series resistance constant.

(1)

Equation (1) is the usual theoretical expression for the diode current.

Also shown in Fig. 1 is an experimental curve. Values of /?, is, and Rs used in calculating the theoretical curve were chosen to make the theoretical curve fit the experi mental data for low currents.

Notice that for diode voltages near 1 V, the diode cur rent drops below the theoretical curve. This means that the higher-power terms in the power series expansion of the current function are smaller at these bias levels than they are at low levels. Rejection of spurious IM responses can therefore be improved by operating the diode with a higher bias voltage than the zero bias usually used.

For still higher voltages, the diode current saturates, and the second-order term in the current function also decreases in magnitude. The second-order term is needed for mixing. Hence there is an optimum bias voltage which results in minimum spurious IM responses and still pro- vi'des adequate mixing.

A second method which will result in the mixer di ode's operating in the region of lower higher-power co efficients is to increase the local oscillator power. Fig. 2 shows the IM characteristics of a single-ended hot-car rier-diode mixer as a function of local-oscillator power. Bias was adjusted for optimum IM rejection at each power level. With 50 milliwatts of local-oscillator power and optimum bias, IM rejection is better than 80 dB. In this experiment, a desired signal was first applied to the mixer and the output observed. Then two spurious sig nals, their frequencies chosen to produce intermodula- tion, were applied to the mixer and their magnitudes adjusted to produce the same output as the desired sig nal (10 dB above the mixer noise, in this case). The two spurious signals had equal magnitudes. Intermodu lation rejection was then computed as the ratio of the spurious input power (one signal) to the desired input power for the same output power.

14

© Copr. 1949-1998 Hewlett-Packard Co.

A. Michael Cowley

M i k e C o w l e y r e c e i v e d h i s B S a n d M S d e g r e e s i n e l e c t r i c a l e n g i n e e r i n g f r o m t h e U n i v e r s i t y o f N o t r e D a m e i n 1 9 5 9 a n d 1 9 6 1 , r e s p e c t i v e l y . I n 1 9 6 5 h e r e c e i v e d t h e P h . D . d e g r e e i n e l e c t r i c a l e n g i n e e r i n g f r o m S t a n f o r d U n i v e r s i t y , w h e r e h e w a s e m p l o y e d a s a r e s e a r c h a s s i s t a n t i n q u a n t u m e l e c t r o n i c s . B e f o r e c o m i n g t o h p a s s o c i a t e s i n 1 9 6 5 , h e t a u g h t e l e c t r i c a l e n g i n e e r i n g f o r a y e a r a t t h e U n i v e r s i t y o f C a l i f o r n i a a t S a n t a B a r b a r a , a n d w o r k e d a s a c o n s u l t a n t i n p h y s i c a l e l e c t r o n i c s .

S i nce j o i n i ng hp assoc ia tes , M i ke has been conce rned m a i n l y w i t h t h e t h e r m a l a n d n o i s e p r o p e r t i e s o f h o t - c a r r i e r d i o d e s . H e h a s p u b l i s h e d s i x p a p e r s i n t h e f i e l d o f s o l i d s t a t e d e v i c e t h e o r y , a n d i s a m e m b e r o f I E E E , T a u B e t a P i , S i g m a X i , a n d t h e A m e r i c a n P h y s i c a l Soc ie ty .

A f t e r r e c e i v i n g h i s B S d e g r e e i n p h y s i c s f r o m t h e U n i v e r s i t y o f N e w H a m p s h i r e i n 1 9 4 3 , J a c k L e p o f f s p e n t t h r e e y e a r s a s a n a v a l o f f i c e r a s s i g n e d t o m i c r o w a v e c o m p o n e n t a n d a n t e n n a d e v e l o p m e n t a t t h e N a v a l R e s e a r c h L a b o r a t o r y . I n 1 9 4 6 h e e n t e r e d C o l u m b i a U n i v e r s i t y a s a g r a d u a t e a s s i s t a n t i n p h y s i c s , a n d r e c e i v e d h i s MA degree in 1948 . F rom 1949 to 1965 he was assoc ia ted w i t h a n u m b e r o f f i r m s a n d g o v e r n m e n t l a b o r a t o r i e s , w o r k i n g o n m i c r o w a v e c o m p o n e n t s , r a d a r , a n t e n n a s , c o m m u n i c a t i o n s s y s t e m s , p a r a m e t r i c a m p l i f i e r s , a n d t u n ne l d i ode amp l i f i e r s .

J a c k j o i n e d h p a s s o c i a t e s i n 1 9 6 5 , a n d a s s u m e d p r o j e c t r e s p o n s i b i l i t y f o r t h e s p u r i o u s r e s p o n s e r e j e c t i o n t e c h n i q u e s p r o j e c t . H e i s n o w s t u d y i n g t e c h n i q u e s f o r c h a r a c t e r i z i n g m i x e r d i o d e s a n d m e t h o d s o f i n t e g r a t i n g s e m i c o n d u c t o r d e v i c e s i n t o m i c r o w a v e c o m p o n e n t s .

Balanced Mixer IM Re jec t ion Even better IM rejection can be obtained in a bal

anced mixer, for reasons which will be analyzed in the next section. Fig. 3 shows the IM characteristics of a balanced hot-carrier diode mixer with more than 100 dB of IM rejection. In measuring the data for Fig. 3, maxi mum input power of 5 mW was used for local-oscillator powers above +18 dBm, and the IM response was below the diode noise level. In the single-ended mixer, the slope of the IM-vs-local-oscillator-power curve is 2, whereas in the balanced mixer the slope is about 20.

Balanced Mixer Analysis In a balanced mixer in which the two diodes are oper

ated at different bias voltages, some of the higher-power coefficients in the power series expansion of one diode's current function can be made to cancel the correspond ing coefficients in the other diode's current function. Cancellation is the reason for the improved IM rejection of the balanced mixer of Fig. 3.

Cancellation of intermodulation is an apparent con tradiction of balanced-mixer theory, which predicts sum mation of this response2. To show how cancellation oc curs, the hypothetical diode characteristic of Fig. 4 was derived. Notice that it falls below the exponential curve just as the typical diode characteristic of Fig. 1 falls below the theoretical (exponential) curve.

 ¡ J . H . L e p o f f , ' S p u r i o u s R e s p o n s e s i n S u p e r h e t e r o d y n e R e c e i v e r s , ' M i c r o w a v e J o u r n a l , V o l . 5 , N o . 6 , J u n e , 1 9 6 2 .

Fig. 4. Hypothetical diode characteristic used in text to illustrate

cancellation of IM-producing higher-power coefficients of diode

characteristics in balanced mixers.

15

© Copr. 1949-1998 Hewlett-Packard Co.

(V)

Fig. 5. Second, fourth, and sixth-power coefficients of hypotheti

cal diode characteristic of Fig. 4. Biasing one diode of balanced

mixer at IV and the other at 2V makes k, 's cancel and kt's = 0,

but changes desired coefficient k, by only a few dB.

: 10

100

I 1 z -

3 0 4 0

TEMPERATURE (°C)

Fig. 6. Temperature dependence of 1M rejection of a

balanced hot-carrier-diode mixer.

In Fig. 5, the coefficients k., k4, and kG for the hypo thetical diode of Fig. 4 are plotted as functions of the bias voltage. Notice that k, = O f or v0 = 1 and for v() = 2, and kc = 0.16 for v,, = 1 and —0.16 for v0 = 2.

This means that if one diode is operated at v.0 = 1 and the other at v0 = 2, the IM response in the mixer output due to the fourth and sixth power terms will be zero. These terms are the most serious causes of intermodu- lation in conventional mixers.

Departure of the diode characteristics from exponen tial at high bias voltages, as shown in both Fig. 1 and Fig. 5, is necessary for cancellation or reduction of IM- producing higher-power terms. Pure exponential behav ior will not produce any benefit from high bias, high local-oscillator power, or unbalanced bias. However, measured characteristics of hot-carrier diodes, point- contact diodes, and other mixing devices do depart from exponential at high bias voltages.

Sensitivity The techniques described here for reducing IM re

sponses would be of little value if they caused an exces sive loss of mixer sensitivity. Fortunately, they do not. For a given local oscillator power and desired-signal input power, the desired response is proportional to the coefficient k¿. This coefficient for the fictitious diode of Fig. 4 is also plotted in Fig. 5. Note that k2 does not change by more than a factor of three for 0 < v0 < 2.5, which means that the sensitivity of a mixer using these diodes would not be changed by more than a few dB by adjustment of the bias for maximum IM rejection.

Effects of Temperature Fig. 6 shows the temperature sensitivity of the IM

characteristic of a balanced hot-carrier-diode mixer. Re jection remains high for a 25 °C range. Operation over wider temperature ranges may be possible if thermistors are used in the bias circuit.

Acknowledgment The authors are grateful to Stewart Krakauer for val

uable suggestions during the work described in this article. • — A. Michael Cowley and Jack H. Lepoff

HEWLETT-PACKARD JOURNAL̂ APRIL Volume 1 8 • Number 8 T E C H N I C A L C A L I F O R N I A F R O M T H E L A B O R A T O R I E S O F T H E H E W L E T T - P A C K A R D C O M P A N Y P U B L I S H E D A T 1 5 0 1 P A G E M I L L R O A D . P A L O A L T O . C A L I F O R N I A 9 4 3 0 4

e d i t o r i a l S t a l l F . J B U R K H A R D . f à ­ . P . D O L A N . L . D S H E R G A L I S . R H . S N Y D E R A f t D i r e c t o r R A f R I C K S O N

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