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1. INTRODUCTION

POWER electronic transformers (PETs) are proposed to replace conventional

transformers and perform voltage regulation and power exchange between generation and

consumption by electrical conversion. The previous researches show that PETs have a great

capacity to receive much more attention due to their merits such as high-frequency link

transformation and flexible regulation of the voltage and power. Although many studies have

been conducted on application and control of PET in power systems, less attention is paid to the

areas of the circuit topologies. The topology of PET can be developed in such a way to achieve

multiport electrical system that converts variable input waveform to the desired output

waveform. In addition, for higher voltage applications or three phase systems, the topology is

expandable as it is modular.

In this paper, a new PET topology named flexible power electronic transformer (FPET)

is proposed. As shown in Fig. 1, it is constructed based on modules and a common dc link,

which is used to transfer energy between ports and isolate all ports from each other. In this

bidirectional topology, each port can be considered as an input or output. Each module consists

of three main parts, including modulator, demodulator, and high frequency isolation transformer

(HFIT). The modulator is a dc– ac converter and the demodulator is an ac–ac converter; both

with bidirectional power flow capability. Each module operates independently and can transfer

power between ports. These ports can have many different characteristics, such as voltage level,

frequency, phase angle, and waveform. As a result, FPET can satisfy almost any kind of

application, which are desired in power electronic conversion systems and meet future needs of

electricity networks. Considering this point, it is named flexible. The simulation results of high-

voltage application are given to clarify the advantages of the proposed FPET over the recently

developed PETs. To show the flexibility of the proposed PET, a prototype is built and tested.

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2. POWER ELECTRONIC TRANSFORMER

An isolated high-frequency link AC/AC converter is often termed as a power electronic

transformer (PET). Because of high frequency operation of the magnetic core they have size and

cost advantages over conventional transformers. These transformers achieve high frequency AC

power transformation without any DC capacitor link. The transformer provides isolation and

voltage transformation while the power converters provide with high frequency operation. Also

termed as high frequency transformers, they have been extensively researched for various

applications, on account of many advantages over conventional line frequency transformers.

The PET has a wide range of applications including electrical distribution systems, wind power

generation etc. In the advantages of PET over conventional transformers are discussed.

2.1 Introduction

In electric power distribution system, transformers perform several functions, such as

voltage transformation, isolation, noise decoupling etc. The conventional distribution

transformers operate at low frequencies (60 Hz) making them bulky and expensive. A power

electronic transformer (PET) operates at much higher frequencies, of the range of several kHz.

The transformer size, which is inversely proportional to the frequency and saturation flux

density, could be reduced under high frequency operation conditions. The PET utilizes power

electronic converters along with a high-frequency transformer to obtain overall size and cost

advantages.

The PET substitutes the conventional 60 Hz transformer at the PCC of a micro-grid,

connecting the later with the utility. This results in an enhanced power management for the

micro-grid as well as decentralized control of the DERs and controllable loads within the micro-

grid. A dynamic control of active and reactive power flow from the utility is possible. It also

allows a bi-directional flow of active power between the utility and the micro-grid. The high

frequency AC power transformation is achieved without a DC link.

Also a smooth transition from grid-connected to isolated mode of micro-grid is possible.

To better acknowledge the claims of the PET, it is essential to understand its operation and

topology, described in the following sections.

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2.2 PET topology

The PET consists of a high frequency transformer with three-phase to single-phase

matrix converters on its primary and secondary as briefly described in chapter 1. Figure.1 shows

a more detailed schematic of the PET. The proposed topology consists of two matrix converters

with high frequency AC link. The primary side of the electronic transformer is supplied by

utility AC source and the secondary side has the equivalent AC source of a micro-grid. The

three phase input AC voltage at the line frequency (60Hz) is first converted into high frequency

(10 kHz) single phase voltage by the input side matrix converter. The output side converter is

also a three phase to single phase matrix converter.

This yields high frequency pulsating single phase AC voltage at the primary and

secondary of the high frequency transformer with leakage reactance Llk . As seen in figure 3-1

the utilities define the limit for the reference power Pref , for a particular micro-grid it serves at

the PCC. The average active power as measured at the output AC source can thus be restricted.

By calculating the corresponding modulation signal for the secondary side matrix converter, the

equivalent phase shift can be achieved. The active power flow between the input side and output

side of the PET is regulated similar to that in a dual active bridge. The active power flow is

controlled by regulating the phase shift between the primary and secondary voltages at the

transformer.

The regulated phase shift angle corresponds to the desired active power limit set by the

utilities. This is achieved by a PI controller designed for the control signal of the PWM strategy

applied for matrix converter modulation. The proportional and integral gain parameters KP and

KI are designed to provide a faster response while eliminating any steady state error.

The PWM strategy also allows a control over the phase shift between the voltage and

current at input as well as output and hence can control the reactive power. Voltage regulation

can thus be achieved by controlling the reactive power at the front-end converter.

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Figure.1 PET schematic

2.3 ISOLATION TRANSFORMER

An isolation transformer is a transformer used to transfer electrical power from a source

of alternating current (AC) power to some equipment or device while isolating the powered

device from the power source, usually for safety. Isolation transformers provide galvanic

isolation and are used to protect against electric shock, to suppress electrical noise in sensitive

devices, or to transfer power between two circuits which must not be connected together.

Suitably designed isolation transformers block interference caused by ground loops. Isolation

transformers with electrostatic shields are used for power supplies for sensitive equipment such

as computers or laboratory instruments.

Strictly speaking any true transformer, whether used to transfer signals or power, is

isolating, as the primary and secondary are not connected by conductors but only by induction.

However, only transformers whose primary purpose is to isolate circuits (opposed to the more

common transformer function of voltage conversion), are routinely described as isolation

transformers. Given this function, a transformer sold for isolation is often built with special

insulation between primary and secondary, and is tested, specified, and marked to withstand a

high voltage between windings, typically in the 1000 to 5000 volt range.

Sometimes the term is exceptionally used to clarify that some transformer, although not

primarily intended for isolation, is a true transformer rather than an autotransformer (whose

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primary and secondary are not isolated from each other).[1] Even step-down power transformers

required, amongst other things, to protect low-voltage equipment from mains voltage by

isolating the secondary and primary such as are used in older "wall warts", are not usually

described specifically as "isolation transformers".

Some very small transformers—e.g. 4 transformers in one tiny dual in-line (DIP) chip

package—used to isolate high-frequency low-voltage (logic) pulse circuits (e.g., 500V RMS

primary–secondary for one second), are described as isolation transformers.

Isolation transformers are commonly designed with careful attention to capacitive

coupling between the two windings. The capacitance between primary and secondary windings

would also couple AC current from the primary to the secondary. A grounded Faraday shield

between the primary and the secondary greatly reduces the coupling of common-mode noise.

This may be another winding or a metal strip surrounding a winding.

2.3.1 Applications

A simple 1. Isolation transformer with an extra dielectric barrier and an electrostatic shield between primary and secondary. The grounded shield prevents capacitive coupling between primary and

secondary windings.

In electronics testing and servicing an isolation transformer is a 1:1 (under load) power

transformer used for safety. Without it, exposed live metal in a device under test is at a

hazardous voltage relative to grounded objects such as a heating radiator or oscilloscope ground

lead (a particular hazard with some old vacuum-tube equipment with live chassis). With the

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transformer, as there is no conductive connection between transformer secondary and earth,

there is no danger in touching a live part of the circuit while another part of the body is earthed.

Electrical isolation is considered to be particularly important on medical equipment, and special

standards apply. Often the system must additionally be designed so that fault conditions do not

interrupt power, but generate a warning.

Isolation transformers are also used for the power supply of devices not at ground

potential. An example is the Austin transformer for the power supply of air-traffic obstacle

warning lamps on radio antenna masts. Without the isolation transformer, the lighting circuits

on the mast would conduct radio-frequency energy to ground through the power supply.

Metal boats are subject to corrosion if they use earthed power from shore when moored,

due to galvanic currents that flow through the water between shore earth and the hull. This can

be avoided by using an isolation transformer with the primary and case connected to shore

earth, and the secondary "floating".

A metal safety screen between primary and secondary is connected to shore earth; in the

event of a fault current in the primary (due, e.g., to insulation breakdown) it will cause the fault

current to return and trip a shore-based circuit breaker rather than making the hull live.

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3. INSULATED GATE BIPOLAR TRANSISTOR (IGBT)

IGBT has been developed by combining into it the best qualities of both BJT and

PMOSFET. Thus an IGBT possesses high input impedance like a PMOSFET and has low on-

state power loss as in a BJT. Further, IGBT is free from second breakdown problem present in

BJT. All these merits have made IGBT very popular amongst power-electronics engineers.

IGBT is also known as metal oxide insulated gate transistor (MOSIGT), conductively-

modulated field effect transistor (COMFET) or gain-modulated FET (GEMFET). It was also

initially called insulated gate transistor (IGT).

The insulated-gate bipolar transistor or IGBT is a three-terminal power

semiconductor device, noted for high efficiency and fast switching. It switches electric power in

many modern appliances: electric cars, variable speed refrigerators, air-conditioners, and even

stereo systems with digital amplifiers. Since it is designed to rapidly turn on and off, amplifiers

that use it often synthesize complex waveforms with pulse width modulation and low-pass

filters.

The IGBT combines the simple gate-drive characteristics of the MOSFETs with the

high-current and low–saturation-voltage capability of bipolar transistors by combining an

isolated-gate FET for the control input, and a bipolar power transistor as a switch, in a single

device. The IGBT is used in medium- to high-power applications such as switched-mode power

supply, traction motor control and induction heating. Large IGBT modules typically consist of

many devices in parallel and can have very high current handling capabilities in the order of

hundreds of amps with blocking voltages of 6,000 V.

The IGBT is a fairly recent invention. The first-generation devices of the 1980s

and early 1990s were relatively slow in switching, and prone to failure through such modes as

latch up and secondary breakdown. Second-generation devices were much improved, and the

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current third-generation ones are even better, with speed rivaling MOSFETs, and excellent

ruggedness and tolerance of over loads.

3.1 BASIC STRUCTURE

Fig illustrates the basic structure of an IGBT. It is constructed virtually in the same

manner as a power MOSFET. There is however , a major difference in the substrate. The n+

layer substrate at the drain in a PMOSFET is now substituted in the IGBT by a p+ layer

substrate called collector C. Like a power MOSFET, an IGBT has also thousands of basic

structure cell connected approximately on a single chip of silicon.

In IGBT, p+ substrate is called injection layer because it injects holes into n -

layer. The n- layer is called drift region. As in other semiconductor devices, thickness of n - layer

determines the voltage blocking capability of IGBT. The p layer is called body of IGBT.The n -

layer in between p+ and p regions serves to accommodate the depletion layer of pn- junction ,

i.e. junction J2.

N-Channel IGBT Cross Section

3.1.1 Equivalent Circuit

An examination of reveals that if we move vertically up from collector to

emitter. We come across p+, n- , p layer s. Thus, IGBT can be thought of as the combination of

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MOSFET and p+ n- p layer s. Thus, IGBT can be thought of as the combination of MOSFET

and p+ n- p transistor Q1 .Here Rd is resistance offered by n – drift region. Approximate

equivalent circuit of an IGBT.

Exact equivalent circuit

The existence of another path from collector to emitter, this path is collector, p +, n-, p (n-

channel), n+ and emitter. There is, thus, another inherent transistor Q2 as n- pn+ in the structure

of IGBT. The interconnection between two transistors Q1 and Q2.This gives the complete

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equivalent circuit of an IGBT. Here Rby is the existence offered by p region to flow of hole

current Ih .

The two transistor equivalent circuit illustrates that an IGBT structure has a

parasitic thyristor in it. Parasitic thyristor is shown in line.

3.2 WORKING

When collector is made positive with respect to emitter, IGBT gets forward biased. With

no voltage between gate and emitter, two junctions between n- region and p region (i.e. junction

J2) are reversed biased; so no current flows from collector to emitter

When gate is made positive with respect to emitter by voltage VG, with gate-

emitter voltage more than the threshold voltage VGET of IGBT, an n-channel or inversion layer,

is formed in the upper part of p region just beneath the gate, as in PMOSFET . This n- channel

short circuits the n- region with n+ emitter regions. Electrons from the n+ emitter begin to flow

to n- drift region through n-channel. As IGBT is forward biased with collector positive and

emitter negative, p+ collector region injects holes into n- drift region .In short; n-drift region is

flooded with electrons from p-body region and holes from p+ collector region. With this, the

injection carrier density in n- drift region increases considerably and as a result, conductivity of

n- region enhances significantly. Therefore, IGBT gets turned on and begins to conducts

forward current IC.

Current Ic , or Ie of two current components:

1. Holes current Ih due to injected holes flowing from collector ,p+ n- p transistor Q1, p-

body region resistance Rby and emitter .

2. Electronic current Ie due to injected electrons flowing from collector, or load, current

IC=emitter current Ie=Ih+Ie.

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Major component of collector current is electronic current Ie, i.e. main current path for

collector, or load, current is through p+, n -, drift resistance Rd and n-channel resistance Rch.

Therefore, the voltage drop in IGBT in its on-state is

Vc e . o n = I c . R c h + I c . Rd + V j i

=voltage drop [in n - channel] + across drift in n- region + across forward

biased p+ n- junction J1.

Here Vji is usually 0.7 to 1v as in a p-n diode. The voltage drop Ic. Rch is due to n-channel

resistance, almost the same as in a PMOSFET. The voltage drop Vdf = Ic.Rd in UGBT is much

less than that in PMOSFET. It is due to substantial increase in the conductivity caused by

injection of electrons and holes in n- drift region. The conductivity increase is the main reason

for the low on-state voltage drop in IGBT than that it is in PMOSFET.

3.3 LATCH-UP IN IGBT

From the above that IGBT structure has two inherent transistors Q1 and Q2,

which constitute a parasitic thyristor. When IGBT is on, the hole current flows through

transistor p+ n- p and p- body resistance Rby. If load current Ic is large, hole component of

current Ih would also be large. This large current would increase the voltage drop Ih. Rby which

may forward bias the base p- emitter n+ junction of transistor Q2. As a consequence, parasitic

transistor Q2 gets turned on which further facilitates in the turn-on of parasitic transistor p+ n- p

labeled Q1. The parasitic thyristor, consisting of Q1 and Q2, eventually latches on through

regenerative action, when sum of their current gains α1+α2 reaches unity as in a conventional

thyristor .With parasitic thyristor on, IGBT latches up and after this, collector emitter current is

no longer under the control of gate terminal. The only way now to turn-off the latched up IGBT

is by forced commutation of current as is done in a conventional thyristor .If this latch up is not

aborted quickly, excessive power dissipation may destroy the IGBT. The latch up discussed

here occurs when the collector current Ice exceeds a certain critical value .the device

manufactures always specify the maximum permissible value of load current Ice that IGBT can

handle without latch up.

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At present, several modifications in the fabrication techniques are listed in the

literatures which are used to avoid latch-up in IGBTs. As such, latch up free IGBTs are

available.

3.4 IGBT Characteristics

The circuit shows the various parameters pertaining to IGBT characteristics.

Static I-V or output characteristics of an IGBT (n-channel type) show the plot of collector

current Ic versus collector-emitter voltage Vce for various values of gate-emitter voltages

VGE1, VGE2 etc .These characteristics are shown below .In the forward direction, the shape of

the output characteristics is similar to that of BJT . But here the controlling parameter is gate-

emitter voltage VGE because IGBT is a voltage controlled device. When the device is off,

junctionJ2 blocks forward voltage and in case reverse voltage appears across collector and

emitter, junction J1 blocks it. Vrm is the maximum reverse breakdown voltage.

The transfer characteristic of an IGBT is a plot of collector current Ic versus gate-

emitter voltage VGE .This characteristics is identical to that of power MOSFET. When VGE is

less than the threshold voltage VGET, IGBT is in the off state.

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Static V-I characteristics

3.5 SWITCHING CHARACTERISTICS

Switching characteristics of an IGBT during turn-on and turn-off are

sketched. The turn-on time is defined as the time between by instance of forward blocking to

forward on-state. Turn-on time is composed of delay time tdn and rise time tr ,i.e. ton=tdn+tr.

The delay time is defined as the time for the collector-emitter voltage to fall from Vce to 0.9

Vce. Here Vce is the initial collector-emitter voltage. Time tdn may also be defined as the time

for the collector current to rise from its initial leakage current Ice to 0.1 Ic. Here Ic is the final

value of the collector current .

The rise time tr is the time during which collector-emitter falls from 0.9VCE to

0.1VCE. IT is also defined as the time for the collector current to rise from 0.1Ic to its final

value Ic. After time ton, the collector current Ic and the collector-emitter voltage falls to small

value called conduction drop=VCES where subscript s denotes saturated value.

The turn-off time is somewhat complex. It consists of three intervals

1. Delay time tdf

2. Initial fall time tf1

3. Final time tf2

I.e. toff=tdf+tf1+tf2

The delay time is the time during which gate voltage falls from VGE to threshold voltage

VGET.As VGE falls to VGET during tdf, the collector current falls from Ic to 0.9 Ic. At the end

of the tdf, collector-emitter voltage begins to rise. The first fall time Tf1 is defined as the time

during which collector current falls from 90 to 20 % of its initial value Ic, or the time during

which collector-emitter voltage rises from Vces to 0.1 Vce.

The final fall time tf2 is the time during which collector current falls from 20 to

10% of Ic, or the time during which collector-emitter voltage rises from 0.1 VCE to final value

VCE.

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3.6 APPLICATIONS OF IGBTS

IGBTs are widely used in medium power applications such as AC and DC

motor drives, UPS systems, power supplies and drives for solenoids, relays and contactors.

Though IGBTs are somewhat more expensive than BJTs, yet they are becoming popular

because of lower gate-drive requirement, lower switching losses and smaller snubber circuit

requirements. IGBT converter are more efficient with less size as well as cost, as compared to

converters based on BJTs. Recently, IGBT inverter induction-motor drives using 15-20KHZ.

Switching frequency favor where audio-noise is objectionable. In most applications, IGBTs will

eventually push out BJTs. At present , the state of the art IGBTs of 1200vots, 500 Amps ratings,

0.25-20 µs turn off time with operating frequency are available.

Comparison of IGBT with MOSFET

Relative merits and demerits of IGBT over PMOSFET are enumerated below.

1. In PMOSFET, the three terminals are called gate, source, drain where as the

corresponding terminal for the IGBTs are gate, emitter and collector.

2. Both IGBT and PMOSFET posses high input impedance.

3. Both are voltage control devices.

4. With rising temperature, increase in on-state resistance in PMOSFET is much

pronounced than in IGBT. So on state voltage drop and losses rise rapidly in

PMOSFET than IGBT, with rising temperature.

5. With rising

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4. PULSE WIDTH MODULATION (PWM)

Pulse Width Modulation (PWM) is the most effective means to achieve constant voltage

battery charging by switching the solar system controller’s power devices. When in PWM

regulation, the current from the solar array tapers according to the battery’s condition and

recharging needs consider a waveform such as this: it is a voltage switching between 0v and

12v. It is fairly obvious that, since the voltage is at 12v for exactly as long as it is at 0v, then a

'suitable device' connected to its output will see the average voltage and think it is being fed 6v -

exactly half of 12v. So by varying the width of the positive pulse - we can vary the 'average'

voltage.

Similarly, if the switches keep the voltage at 12 for 3 times as long as at 0v, the average

will be 3/4 of 12v - or 9v, as shown below and if the output pulse of 12v lasts only 25% of the

overall time, then the average is

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By varying - or 'modulating' - the time that the output is at 12v (i.e. the width of the

positive pulse) we can alter the average voltage. So we are doing 'pulse width modulation'. I

said earlier that the output had to feed 'a suitable device'. A radio would not work from this: the

radio would see 12v then 0v, and would probably not work properly. However a device such as

a motor will respond to the average, so PWM is a natural for motor control.

4.1 PULSE WIDTH MODULATOR

So, how do we generate a PWM waveform? It's actually very easy, there are circuits

available in the TEC site. First you generate a triangle waveform as shown in the diagram

below. You compare this with a d.c voltage, which you adjust to control the ratio of on to off

time that you require. When the triangle is above the 'demand' voltage, the output goes high.

When the triangle is below the demand voltage, the

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When the demand speed it in the middle (A) you get a 50:50 output, as in black. Half the

time the output is high and half the time it is low. Fortunately, there is an IC (Integrated circuit)

called a comparator: these come usually 4 sections in a single package. One can be used as the

oscillator to produce the triangular waveform and another to do the comparing, so a complete

oscillator and modulator can be done with half an IC and maybe 7 other bits.

The triangle waveform, which has approximately equal rise and fall slopes, is one of the

commonest used, but you can use a saw tooth (where the voltage falls quickly and rinses

slowly). You could use other waveforms and the exact linearity (how good the rise and fall are)

is not too important.

Traditional solenoid driver electronics rely on linear control, which is the application of

a constant voltage across a resistance to produce an output current that is directly proportional to

the voltage. Feedback can be used to achieve an output that matches exactly the control signal.

However, this scheme dissipates a lot of power as heat, and it is therefore very inefficient.

A more efficient technique employs pulse width modulation (PWM) to produce the

constant current through the coil. A PWM signal is not constant. Rather, the signal is on for part

of its period, and off for the rest. The duty cycle, D, refers to the percentage of the period for

which the signal is on. The duty cycle can be anywhere from 0, the signal is always off, to 1,

where the signal is constantly on. A 50% D results in a perfect square wave. (Figure 1)

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A solenoid is a length of wire wound in a coil. Because of this configuration, the

solenoid has, in addition to its resistance, R, a certain inductance, L. When a voltage, V, is

applied across an inductive element, the current, I, produced in that element does not jump up to

its constant value, but gradually rises to its maximum over a period of time called the rise time

(Figure 2). Conversely, I does not disappear instantaneously, even if V is removed abruptly, but

decreases back to zero in the same amount of time as the rise time.

Therefore, when a low frequency PWM voltage is applied across a solenoid, the current

through it will be increasing and decreasing as V turns on and off. If D is shorter than the rise

time, I will never achieve its maximum value, and will be discontinuous since it will go back to

zero during V’s off period (Figure 3).* In contrast, if D is larger than the rise time, I will never

fall back to zero, so it will be continuous, and have a DC average value. The current will not be

constant, however, but will have a ripple.

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At high frequencies, V turns on and off very quickly, regardless of D, such that the

current does not have time to decrease very far before the voltage is turned back on. The

resulting current through the solenoid is therefore considered to be constant. By adjusting the D,

the amount of output current can be controlled. With a small D, the current will not have much

time to rise before the high frequency PWM voltage takes effect and the current stays constant.

With a large D, the current will be able to rise higher before it becomes constant.

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4.2 WHY THE PWM FREQUENCY IS IMPORTANT

The PWM is a large amplitude digital signal that swings from one voltage extreme to the

other. And, this wide voltage swing takes a lot of filtering to smooth out. When the PWM

frequency is close to the frequency of the waveform that you are generating, then any PWM

filter will also smooth out your generated waveform and drastically reduce its amplitude. So, a

good rule of thumb is to keep the PWM frequency much higher than the frequency of any

waveform you generate.

Finally, filtering pulses is not just about the pulse frequency but about the duty cycle

and how much energy is in the pulse. The same filter will do better on a low or high duty cycle

pulse compared to a 50% duty cycle pulse. Because the wider pulse has more time to integrate

to a stable filter voltage and the smaller pulse has less time to disturb it the inspiration was a

request to control the speed of a large positive displacement fuel pump. The pump was sized to

allow full power of a boosted engine in excess of 600 Hp.

At idle or highway cruise, this same engine needs far less fuel yet the pump still

normally supplies the same amount of fuel. As a result the fuel gets recycled back to the fuel

tank, unnecessarily heating the fuel. This PWM controller circuit is intended to run the pump at

a low speed setting during low power and allow full pump speed when needed at high engine

power levels.

4.3 MOTOR SPEED CONTROL (POWER CONTROL)

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Typically when most of us think about controlling the speed of a DC motor we think of

varying the voltage to the motor. This is normally done with a variable resistor and provides a

limited useful range of operation. The operational range is limited for most applications

primarily because torque drops off faster than the voltage drops.

Most DC motors cannot effectively operate with a very low voltage. This method also

causes overheating of the coils and eventual failure of the motor if operated too slowly. Of

course, DC motors have had speed controllers based on varying voltage for years, but the range

of low speed operation had to stay above the failure zone described above.

Additionally, the controlling resistors are large and dissipate a large percentage of

energy in the form of heat. With the advent of solid state electronics in the 1950’s and 1960’s

and this technology becoming very affordable in the 1970’s & 80’s the use of pulse width

modulation (PWM) became much more practical. The basic concept is to keep the voltage at the

full value and simply vary the amount of time the voltage is applied to the motor windings.

Most PWM circuits use large transistors to simply allow power On & Off, like a very fast

switch.

This sends a steady frequency of pulses into the motor windings. When full power is

needed one pulse ends just as the next pulse begins, 100% modulation. At lower power settings

the pulses are of shorter duration. When the pulse is On as long as it is Off, the motor is

operating at 50% modulation. Several advantages of PWM are efficiency, wider operational

range and longer lived motors. All of these advantages result from keeping the voltage at full

scale resulting in current being limited to a safe limit for the windings.

PWM allows a very linear response in motor torque even down to low PWM% without

causing damage to the motor. Most motor manufacturers recommend PWM control rather than

the older voltage control method. PWM controllers can be operated at a wide range of

frequencies. In theory very high frequencies (greater than 20 kHz) will be less efficient than

lower frequencies (as low as 100 Hz) because of switching losses.

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The large transistors used for this On/Off activity have resistance when flowing current,

a loss that exists at any frequency. These transistors also have a loss every time they “turn on”

and every time they “turn off”. So at very high frequencies, the “turn on/off” losses become

much more significant. For our purposes the circuit as designed is running at 526 Hz. Somewhat

of an arbitrary frequency, it works fine.

Depending on the motor used, there can be a hum from the motor at lower PWM%. If

objectionable the frequency can be changed to a much higher frequency above our normal

hearing level (>20,000Hz).

4.4 PWM CONTROLLER FEATURES

This controller offers a basic “Hi Speed” and “Low Speed” setting and has the option to

use a “Progressive” increase between Low and Hi speed. Low Speed is set with a trim pot inside

the controller box. Normally when installing the controller, this speed will be set depending on

the minimum speed/load needed for the motor. Normally the controller keeps the motor at this

Lo Speed except when Progressive is used and when Hi Speed is commanded (see below). Low

Speed can vary anywhere from 0% PWM to 100%.

Progressive control is commanded by a 0-5 volt input signal. This starts to increase

PWM% from the low speed setting as the 0-5 volt signal climbs. This signal can be generated

from a throttle position sensor, a Mass Air Flow sensor, a Manifold Absolute Pressure sensor or

any other way the user wants to create a 0-5 volt signal. This function could be set to increase

fuel pump power as turbo boost starts to climb (MAP sensor). Or, if controlling a water

injection pump, Low Speed could be set at zero PWM% and as the TPS signal climbs it could

increase PWM%, effectively increasing water flow to the engine as engine load increases.

This controller could even be used as a secondary injector driver (several injectors could

be driven in a batch mode, hi impedance only), with Progressive control (0-100%) you could

control their output for fuel or water with the 0-5 volt signal.

Progressive control adds enormous flexibility to the use of this controller. Hi Speed is

that same as hard wiring the motor to a steady 12 volt DC source. The controller is providing

100% PWM, steady 12 volt DC power. Hi Speed is selected three different ways on this

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controller: 1) Hi Speed is automatically selected for about one second when power goes on.

This gives the motor full torque at the start. If needed this time can be increased ( the value of

C1 would need to be increased). 2) High Speed can also be selected by applying 12 volts to the

High Speed signal wire. This gives Hi Speed regardless of the Progressive signal.

When the Progressive signal gets to approximately 4.5 volts, the circuit achieves 100%

PWM – Hi Speed.

4.5 HOW DOES THIS TECHNOLOGY HELP

The benefits noted above are technology driven. The more important question is how the

PWM technology jumping from a 1970’s technology into the new millennium offers:

• LONGER BATTERY LIFE

– reducing the costs of the solar system

– reducing battery disposal problems

• MORE BATTERY RESERVE CAPACITY

– increasing the reliability of the solar system

– reducing load disconnects

– Opportunity to reduce battery size to lower the system cost

• GREATER USER SATISFACTION:

– get more power when you need it for less money

5. MODELLING OF CASE STUDY

5.1 PROPOSED POWER CIRCUIT OF FPET

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The proposed circuit is shown in Fig. 2. It should be mentioned that the proposed

topology can be expanded by connecting modules in series or parallel to obtain higher voltage

or current ratings, and to form star/delta connections for three phase applications.

As shown in Fig. 2(a), each port is composed of a full bridge dc-link inverter (FBDCI),

HFIT, and a Cyclo-converter. This topology consists of independent and similar modules and

each port can work independently. Thus, the analysis of one port is sufficient to introduce whole

topology. The FBDCI (modulator) can operate as an inverter when it converts the dc-link

voltage to an ac waveform at the HFIT side. It can operate as an active rectifier when it converts

the ac waveform of the HFIT to the dc-link voltage. The FBDCI is used to achieve zero-voltage

level, adjustable pulse width, and symmetrical switching. In addition, the number of switches

can be reduced to obtain simpler circuit than the latter, shown in Fig. 2(b). In this case, one of

the half-bridge circuits can be considered as the reference or master leg. Once gate pulses for

the master leg (i.e., switches and ) are provided, the gate pulses of the other legs (slave legs)

have a phase shift respect to the master leg. Using this control strategy, the number of switches

can be reduced to half.

Fig2.Proposed circuit of the FPET (a) Basic topology and (b) reduced switch topology

The modulator can be described as follows:

1) Bi directional power flow capability;

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2) Adjustable switching frequency that feet voltage pulses frequency into the pass band of

HFIT;

3) Stored energy in the dc link (if the modulator is in active rectifier mode). For

cycloconverters, several circuit topologies can be proposed using unidirectional or bidirectional

switches.

In this paper, a typical cycloconverter with two bidirectional switches operates as the

demodulator. The demodulator converts high frequency voltage (i.e.) to low frequency voltage

(i.e., Vpr1 ) and vice versa. The specifications of the demodulator are listed as follows:

1) Bidirectional power flow capability; and

2) Providing zero voltage switching by turning the switches of cycloconverter ON/OFF, while

voltage of HFIT riches to zero.

5.2 MODULATION AND DEMODULATION OPERATION PRINCIPLES

The well-known phase shift modulation (PSM) method is shown in Fig. 3. The

definition of parameters is given in Table I.

TABLE IDEFINITION OF PARAMETERS

The voltage regulation is performed by the FBDCI using PSM method. The cycloconverter

chooses the PSM pulses in such a way to provide positive or negative voltage polarity at the

output. In this figure, the cycloconverter provides positive output voltage polarity as an

example. On one hand, the switches of cycloconverter turn ON/OFF with a time delay (Tcd )

respect to those of FBDCI, so they operate under zero voltage condition.

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On the other hand, the switches have a small overlapping time to provide a path for Lf

current to avoid high stresses at switching instants. Thus, the switches operate at soft switching

condition. The leakage inductance of HFIT should be minimized as much as possible. In

practice, snubber circuits must be used to damp the stored energy in the leakage inductance of

HFIT

Fig.3. Principle of PSM method

According to Fig. 3, the duty cycle of FBDCI is defined as follows:

The modulated voltage at the secondary side for one duty cycle is expressed by (2)

The modulated voltage at the output of cycloconverter (Vc) is determined as follows:

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Where sign (tk ) function determines the polarity of Vc that can be positive or negative

according to the desired output voltage and presented by (4), as shown at the bottom of the

page.

A. PSM Control Circuit

Fig. 4. Schematic presentation of PSM controller.

The control circuit is responsible for providing pulse gate of dc link switches and the

cycloconverter. The implementation of PSM is shown in Fig. 4. The input data address consists

of four lines. The first line is polarity of output voltage signi. The second line is switch-enabled

of cycloconverter (EnableCi ). The third line is switch-enabled of dc link (EnableSi ). The

fourth line provides the duty cycle data of the ith port. The enabled lines are provided by the

startup and protection circuits.

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B. Utilization of Ports as a Voltage Source

As an example when a port (assuming ith port) is designed to operate as a voltage

source, it can provide a constant voltage regardless of the active or reactive power that is

exchange between the port and the grid. So, a controllable voltage at the output of

cycloconverter can be obtained and it is given by

Where (t) is the reference voltage. According to (4), one may obtain the following

approximation:

Where the asterisk symbols show the next stage values Therefore, the duty cycle and the

sign function are achieved as follows:

Because of high switching frequency, it is expected to assume is constant over time

period of kTs < t < (k + 1)Ts. The duty cycle is a function of dc-link voltage (Vd (kTs)) and the

turn winding of the HFIT at the ith port. The block diagram of controller is shown in Fig. 5.

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5.3 ENERGY BALANCE IN FPET

In every system, there is a balance among losses, input energy and output energy. This

balance for FPET is presented as follows

Fig.5. Control circuit of a typical port that operates as a voltage source

TABLE II

DESCRIPTION OF PARAMETERS PREARRANGED IN (10)

Where Wi, WCd, and Wloss are the input/output energy, stored energy at dc link and

losses, respectively. Neglecting the power losses, (8) can be approximated by

To achieve power equilibrium in Cd and have constant dclink voltage, some of the ports

should absorb and inject desired active power. The algorithm for regulation of dc-link voltage is

as follows:

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Step 1: At the start-up instant, following two methods can be used to charge the dc-link

capacitor to the desired value.

1) The dc-link capacitor can be charged by an extra dc source. As the desired dc-link voltage

achieved, the dc source should be disconnected.

2) The cycloconverter can provide a high frequency voltage across HFIT. When the voltage

passes through HFIT, it changes to a dc voltage across dc-link capacitor by the body diodes of

FBDCI switches. The dc voltage can charge the capacitor considering the winding ratio of

HFIT. The startup current is limited by Lf .

Step 2: dc-link voltage checking.

1. If then there is no need for adjustments. The

ΔVd, Ref is a fraction of Vd, Ref that is required to provide Hystersis band.

2. If then voltage should be

regulated and the port powers should be adjusted.

Step 3: Return to the second step.

A. Balancing Ports:

For another solution to regulate voltage of dc link, some ports are considered as

“balancing ports” that provide energy to balance dc-link voltage in FPET. One of the main

objectives of these kinds of ports is to control voltage level in the dc-link voltage, particularly

when over voltage or voltage drop occurs in the dc link. Assuming the ith port is chosen as the

balancing port, the main component of the cycloconverter voltage, and output of the port are

given as follows:

Fig.6. Simplified diagram of FPET

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The definition of the parameters is given in the Table II. Therefore, neglecting the

resistance of output filter inductance, the active power of the port is obtained as follows:

Applying the differences between Vd and Vd,Ref as an error signal to a typical PI

controller, the value of required Pi can be estimated. According to (6) and (7), the duty cycles

are achieved.

5.4 DESIGN PROCEDURE

A. DC-Link Capacitor

Fig. 6 shows the voltage and currents of all ports and the dc link capacitor. The

following equation presents the instantaneous power balance of the losses in FPET.

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Fig.7. Proposed HV FPET

The voltage and current of ports can have different polarity and directions. If the

currents and voltages of ports have sinusoidal waveforms, then (12) can be rewritten as follows:

Now, the input power of dc link can be expressed as follows:

(14)

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This input power consists of two components. The first component is the pulsation

power with angular frequency of 2ωi and the second one is the dc power . Assuming

Vd,ref as the voltage of capacitor and Id as the average current, (14) can be rewritten as follows:

The ripple voltage of the dc-link capacitor (ΔVd ) can be approximated as follows:

Thus, the minimum value of Cd can be calculated for the maximum voltage ripple.

B. Reference Voltage of DC Link and Winding Ratio of HFIT

From practical point of view, lower dc-link voltage results in lower voltage stress of

switches. But according to (17), as Vd, Ref decreases, the voltage ripple increases. In addition,

the decrease of the dc-link voltage increases the current of dc link switches. Consequently, by

selecting an appropriate dc link reference voltage (Vd,Ref ) and the maximum ripple voltage,

the minimum dc-link voltage (Vd,min) can be determined. In the worst condition, the lowest dc-

link voltage (Vd,min), maximum duty cycle (D = Dmax) and the maximum magnitude of

desired.

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TABLE III

PARAMETERS OF PETS

Fig.8. Port voltage and current of HV FPET

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Fig.9. Load voltage and current of three-phase output

Voltage (Vi,max) can determine the winding ratio as follows:

C. Matching Inductance Lf

Matching inductance Lf should limit the output current to its maximum acceptable value

(Ii,max) during the switching period (Ts). For the ith port, the following assumptions can be

considered:

Where Δifi is the variation of the cycloconverter current for one switching period Based

on these assumptions, Lf is determined by

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5.5 COMPARISON STUDY

A comparison study is given to clarify the advantageous and disadvantageous of the

FPET. A three-phase system, contains six ports, is compared to the similar PETs. First, some of

the pros and cons of bidirectional FPET in comparison to the unidirectional topologies should

discuss. In the unidirectional systems, input power factor is not controllable but in bidirectional

structures input or even output power factor can be adjusted. This means that the reactive and

active power of each port can be regulated. Also for DG systems like wind turbine, bidirectional

capability is indispensable. Energy management for energy efficient systems is another

application of this feature. A detail comparison study (e.g., cost, efficiency, quality, etc.) is

given in Table V to clarify the pros and cons of FPET and the existing topologies proposed.

As can be seen from Table V, conversion efficiency of FPET is relatively low in

comparison to the similar circuits topologies proposed. The main reason is the usage of power

snubber, and voltage clamp circuits, which damp absorbed energy in leakage inductors of HFIT.

To reduce the size of protection circuits in FPET, a PSM approach is utilized, so the

cycloconverter switches just select the PSM pulses and can commutate naturally. Therefore, the

switches communicate at almost zero voltage. In addition, because of overlap technique the

voltage surge is reduced over the switches and the continuous current flow in the output filter

(Lf) is not interrupted.

In addition, Table V shows some of the most noticeable applications of FPET. Dynamic

voltage restorer (DVR) and active filter (AF) applications can be satisfied by the FPET, because

it can connect to the grid in series or/and in parallel. Desired voltage and current can provide by

the flexibility of FPET in providing various waveforms (see Section VI). FPET can provide

desired waveform in each phase (or port) independently, so this can be used in universal power

quality conditioner (UPQC). FPET can transfer active and reactive power from one port or

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phase to another port or one phase. This in power distribution system is very useful for interline

power flow controller (IPFC).

Additionally, FPET can provide symmetrical three-phase voltage from an asymmetrical ac

source in the form of an uninterruptible power supply application (UPS). FPET can play a role

in providing useful power from variable low-voltage dc sources. That is suitable for renewable

energy applications such as photovoltaic and fuel cell. Design simplicity and expandability (to

achieve higher ratings) are other advantageous of FPET.

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6. CASE STUDY AND RESULTS

Flexible Transformer

Fig 6.1 flexible power electronic transformer circuit diagram 1

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Fig 6.2 Output voltage at port 8

Fig 6.3 dc link voltage

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Fig 6.4 load voltages of three phase output at ports 6,7,8

Fig 6.5 load voltage at port 1

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Fig 6.6 load current at port 1

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Flexible Transformer 1

Fig 6.7 flexible power electronic transformer circuit 1

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Fig 6.8 load voltage at port 8

Fig 6.9 dc link voltage

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Fig 6.10 load current of three phase output at ports 6,7,8

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CONCLUSION

Based on the requirement of a flexible power conversion system, FPET is proposed to

facilitate many requirements that are expected in power electronic and distribution systems. The

proposed topology is flexible enough to provide bidirectional power flow and has as many ports

as it is required. For low-voltage application, FPET can correct power factor and can adjust the

waveform and frequency of the output voltage. The proposed topology can be expanded for

high voltage and high current applications. The dc link plays a significant role to provide energy

balance, power management in the circuit and independent operation of ports. The measurement

results verify the basic theoretical concepts of this paper. The advantages of the FPET are:

bidirectional power flow capability of ports, module-based topology, which can be used in

different forms, independent operation of ports, flexibility in power amount and direction in all

ports, and double galvanic isolation between each port, as well as using only one storage

element.

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[2] S. H. Hosseini, M. Sabahi, and A. Y. Goharrizi, “Multi-function zero voltage and zero-

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