Post on 10-Feb-2021
Hindawi Publishing CorporationActive and Passive Electronic ComponentsVolume 2013, Article ID 953498, 10 pageshttp://dx.doi.org/10.1155/2013/953498
Research ArticleA Low-Power Ultrawideband Low-Noise Amplifier in0.18 πm CMOS Technology
Jun-Da Chen
National Quemoy University, Department of Electronic Engineering, University Road, Jinning Township, Kinmen 892, Taiwan
Correspondence should be addressed to Jun-Da Chen; chenjd@nqu.edu.tw
Received 11 July 2013; Revised 17 October 2013; Accepted 4 November 2013
Academic Editor: Rezaul Hasan
Copyright Β© 2013 Jun-Da Chen.This is an open access article distributed under the Creative Commons Attribution License, whichpermits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.
This paper presents an ultrawideband low-noise amplifier chip using TSMC 0.18πm CMOS technology. We propose a UWB lownoise amplifier (LNA) for low-voltage and low-power application. The present UWB LNA leads to a better performance in termsof isolation, chip size, and power consumption for low supply voltage. This UWB LNA is designed based on a current-reusedtopology, and a simplified RLC circuit is used to achieve the input broadband matching. Output impedance introduces the LCmatching method to reduce power consumption. The measured results of the proposed LNA show an average power gain (S
21) of
9 dB with the 3 dB band from 3 to 5.6GHz. The input reflection coefficient (S11) less than β9 dB is from 3 to 11 GHz. The output
reflection coefficient (S22) less than β8 dB is from 3 to 7.5 GHz. The noise figure 4.6β5.3 dB is from 3 to 5.6GHz. Input third-order-
intercept point (IIP3) of 2 dBm is at 5.3 GHz. The dc power consumption of this LNA is 9mW under the supply of a 1 V supply
voltage. The chip size of the CMOS UWB LNA is 1.03 Γ 0.78mm2 in total.
1. Introduction
The ultrawideband (UWB) system has become one of themajor technologies for wireless communication systemsand local area networks. The IEEE 802.15.3a ultrawideband(UWB) system uses a specific frequency band (3.1 GHzβΌ10.6GHz) to access data and employs the system of orthogo-nal frequency-divisionmultiplexing (OFDM)modulation [1β3].The frequency band consists of four groups: A, B, C, andD,with thirteen channels. Each channel bandwidth is 528MHz,as shown in Figure 1.The system operates across a wide rangeof frequency 3.1β5GHz or 3.1β10.6GHz. The low frequencyband from 3.1 to 5GHz has been allocated for developingthe first generation of UWB systems [4]. The system hasseveral advantages such as low complexity, low cost, and ahigh data rate for the wireless system. In the front-end systemdesign, a low-noise amplifier (LNA) is the first block in thereceiver path of a communication system.The chief objectiveof the LNA is to reach the low-noise figure to improve theoverall system noise [5]. The LNA must minimize the noisefigure over the entire bandwidth, feature flat gain, goodlinearity, wideband input-output matching, and low power
consumption. In the radio frequency circuit design,GaAs andbipolar transistors have performed fairly well. Nevertheless,these processes lead to increased cost and greater complexity.The RF front-end circuits using CMOS technology canprovide a single-chip solution, which greatly reduces the cost[6]. With the rapid improvement of CMOS technology, itcan implement RF ICs with CMOS. Three major topologiesof the present wideband low-noise amplifiers are used. First,distributed amplifiers [7β9] can provide good linearity andwidebandmatching but require high power consumption anda large chip area because of the multiple amplifying stages.Second, resistive shunt feedback amplifiers [4, 10] providegoodwidebandmatching andflat gain but require high powerconsumption. Last, Chebyshev filter amplifiers [11, 12] adopt awideband LC bandpass filter for input matching and a sourcefollower for outputmatching.This type of wideband amplifierdissipates a small amount of dc power. The Chebyshevfilter matching design requires three inductors. Therefore, avery large chip area is required. This paper proposes RLCbroadband matching for LNA design. The primary goal isto obtain wideband input-output matching and to reducethe chip area and power consumption. The LNA topology is
2 Active and Passive Electronic Components
proposed in Section 2. The complete analysis of the designmethodology of the widebandmatching network is presentedin Section 3. In Section 4, implementation and measurementresults are presented. The conclusion is presented in the lastsection.
2. Low-Voltage WidebandCascode LNA Design
Figure 2 shows the basic cascode LNA. The current-reusedconfiguration can be considered as two common-sourceamplifiers (π
1andπ
2).The current shared cascode amplifier
provides a low-power characteristic under the low voltagesupply. Because π
1and π
2share the same bias current,
the total power consumption is minimized [6, 12, 16]. For acascode amplifier, the overall noise figure can be obtainedin terms of the noise figure (NF) and gain of each stage asfollows:
πΉtotal = πΉ1 +πΉ2β 1
πΊπ΄1
, (1)
where πΉtotal is the total NF and πΉ1 and πΉ2 are the NF values ofthe first and second stage; respectively. πΊ
π΄1is the power gain
of this first stage, therefore, in the LNA circuits, the noise ofthe first stage is more critical. Consequently, the gain of thetransistorβs π
1must be high enough to suppress the noise.
The dominant noise sources for active MOSFET transistorsare flicker and thermal noises.The flicker noise is modeled asthe voltage source in series with the gate of value:
π2π(π) =
πΎ
ππΏπΆππ₯π, (2)
where the constant πΎ is dependent on device characteristicsand can vary widely for different devices in the same process.The variables π, πΏ, and πΆ
ππ₯represent the width, length,
and gate capacitance per unit area, respectively. The flickernoise is inversely proportional to the transistor area,ππΏ. Inother words, a larger device reduces this noise. The noiseprocess of thermal noise is random. The MOSFET thermalnoise includes threemain noise sources, as shown in Figure 3:distributed gate resistance noise (π2rg), drain current noise(π2nd), and gate current noise (π
2
nd).The thermal noise source mainly comes from the drain
current noise and gate-induced noise of the transistors.Multifinger layout technology is applied to reduce the gate-induced noise. According to the MOSFET noise analysis in[17, 18], we must express the noise figure in a way that explic-itly considers power consumption. In order to determine thewidth ofπ
1and find out the best NF, we use the dependency
among noise factor (πΉ), power dissipation (PD), and qualityfactor (π
πΏ) to find out the optimal value [17]. Based on
these parameters, the related curves, NF (dB) versus ππΏ, are
plotted in Figure 4 by MATLAB simulation software. Thereis a compromise between noise performance and power gainin Figure 4. Therefore, the π
πΏcan be determined to be 4, as
PD equals to 10mW. The optimal width of MOSFET can beobtained by [17, 18]:
πopt =3
2
1
πππΏπΆππ₯π πππΏ
β1
3πππΏπΆππ₯π π
, (3)
where ππis the operation frequency, πΏ is the gate length, and
πΆox is the gate capacitance per unit area. Equation (3) showsthat the optimal width for π
1is 240πm, for πo = 3.4GHz,
πΏ = 0.18 πm, πΆππ₯= 8.62 F/m2, π
π= 50Ξ©, and π
πΏ= 4.
Figure 5 shows the proposed UWB LNA schematic.The current sharing cascode amplifier provides low-powercharacteristics with low voltage supply. Note that π
2is
the common-gate stage of the cascode configuration, whicheliminates theMiller effect and provides better isolation fromthe output return signal [18, 19]. The passive components πΆ
1,
πΆ2, πΆ3, π 1, and πΏ
1are adopted for the matching network
at the input to resonate over the entire frequency band. Theresistors π
2and π
πare used to provide bias voltage for the
transistors π1and π
2. The capacitor πΆ
4provides signal
coupling between the two stages. The capacitor πΆ5bypasses
the ac current to the ground and avoids coupling to the firststage. This affects gain flatness; therefore, it is possible toprovide an ideal ac ground, but gain flatness is not affectedin the design. πΏ
2is the inductor load of the first stage. The
output matching network is composed of πΏ3, πΏ4, πΆ6, and πΆ
7.
In the cascode stages, the first stageβs noise figure contributionis more than the second stageβs. The first stageβs transistorsize and bias point should be optimized for the low NF [17].The second stageβs transistor size and bias point should beoptimized for high linearity. The cascode LNA design is fullof trade-offs between optimal gain, low noise figure, inputand output matching, linearity, and power consumption.Moreover, to avoid oscillation, the parasitic couplings haveto be minimized [16]. The sizes of the devices of the LNA areshown in Table 1.
3. Proposed Wideband MatchingNetwork Analysis
3.1. Input Impedance Matching. As shown in Figure 2, in theconventional narrow band LNA design, the input impedanceof the input stage (π
1) can be written as
πin = π (πΏπ + πΏπ) +1
π πΆgs1+ (
ππ1
πΆgs1)πΏπ , (4)
where πΆgs1 is the gate-source capacitance and ππ1 is thetransconductance of the input transistorπ
1. We choose ap-
propriate values of inductance (πΏπ+πΏπ ) and capacitanceπΆgs1,
which resonate at a certain frequency. The real term can bemade equal to 50Ξ©. Equation (3) determines the width ofπ
1
and finds the best NF. For wideband design, it is difficult tolet the imaginary part of (4) remain zero for a wide range.To discuss UWB input impedance matching, we need toconsider the standard form of the second-order filter:
π (π) =π2π2+ π1π + ππ
π2 + π (ππ/πin) + π
2
π
=π2π2+ π1π + ππ
π2 + πB + π2π
, (5)
Active and Passive Electronic Components 3
3432 3960 4488(MHz)
5016 5808 6336 6864 7392 7920 8448 8976 9504 10032
Group A Group B Group C Group D
Figure 1: The IEEE 802.15.3a spectrum.
Table 1: Device sizes of the proposed UWB LNA.
Device πΆ1(pF)πΆ2
(pF)πΏ1
(nH)πΆ3
(pF)π 1
(kΞ©)π π
(kΞ©)πΆ4
(pF)πΏ2
(nH)πΆ5
(pF)π 2
(kΞ©)πΏ3
(nH)πΏ4
(nH)πΆ6
(pF)πΆ7
(pF)π1-π2
(πm)Size 9.51 0.11 0.91 7.6 0.14 3.84 5.7 5.53 5.7 3.84 1.19 0.59 0.16 1.23 240/0.18
M2
M1
Ls
Lg
Vbias
Zin
Figure 2: Cascode low noise amplifier.
V2rg
gsi2ng i
2nd
Cgs rogmV
Rg
Figure 3: CMOS noise model.
where π2, π1, and π
πare numerator coefficients determining
the type of second-order filter function. ππis called the pole
frequency,πin is termed the pole factor, and π΅ is called band-width. According to π΅ = π
π/πin, the bandwidth is inversely
proportional to the πin if the value of the pole frequency ππis fixed.
0 1 2 3 4 5 6 7 80.0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
NF
(dB)
PD = 5 mWPD = 10 mW
PD = 15 mWPD = 20 mW
QL
Figure 4: Noise Figure versus ππΏfor several power dissipations at
3.4GHz.
R2
L3 L4 C7
C6
C5
C4L2
M2
M1
C3
R1C2
C1 L1
RgVbias
RFin
RFout
VDD
Figure 5: Schematic of the proposed UWB LNA.
4 Active and Passive Electronic Components
Rs = 50Ξ©
Vs
Zin
C1
C2
C3
R1
L1Cgs1
Figure 6: The proposed wideband matchingapproximate first stageinput matching small-signal equivalent circuit.
Figure 6 shows the proposed wideband matching and asmall-signal equivalent circuit in the first stage. The inputimpedance of the RLC network can be written as
πin =1
ππΆ1
+1
ππΆ2
// [ππΏ1+ (π 1+
1
ππΆ3
) //1
ππΆgs1]
= π in + ππin.
(6)
Because πΆ3capacitance value is much larger than πΆgs(πΆ3 β«
πΆgs), the (π 1 + (1/ππΆ3))//(1/ππΆgs1) will approximate (π 1 +(1/ππΆ
3)):
πin =1
ππΆ1
+1
ππΆ2
// (ππΏ1+ (π 1+
1
ππΆ3
)) = π in + ππin,
πin β(π + (π
1/πΏ1)) ((πΆ
1+ πΆ2) /πΆ1πΆ2)
π2 + π (π 1/πΏ1) + ((πΆ
2+ πΆ3) /πΏ1(πΆ2πΆ3)),
(7)
where πΆgs1 is the gate-source capacitance ofπ1. This equiv-alent circuit can roughly be an RLC second-order filterstructure. In (7), the quality factor of the filter circuit can begiven by
ππβ
1
2πβ
πΆ2+ πΆ3
πΏ1(πΆ2πΆ3), (8)
π β1
π 1
βπΏ1(πΆ2+ πΆ3)
πΆ2πΆ3
, (9)
where ππis the resonant frequency. In (8) and (9), to obtain
broader bandwidth, a low-quality factor needs to be added tothe passive devices. The narrowband LNA can be convertedinto a wideband amplifier by properly selecting passivedevices. According to (8), the capacitors πΆ
2, πΆ3and inductor
πΏ1will be optimized at the resonant frequency. We use the
CAD of Agilent ADS to analyze the circuit performanceof input impedance matching. According to (8) and (9), π
π
is inversely proportional to πΏ1while π is proportional to
πΏ1. Therefore, bandwidth is inversely proportional to πΏ
1.
Figure 7 is fixed as πΆ1(9.51 pF), πΆ
2(0.11 pF), πΆ
3(7.6 pF), and
π 1(140Ξ©), showing that the resonant frequency is inversely
proportional to the πΏ1. Bandwidth is inversely proportional
3 4 5 6 7 8 9 102 11Frequency (GHz)
β20
β25
β15
β10
β5
0
L1 = 1.3nHL1 = 1.15nHL1 = 0.91 nH
S 11
(dB)
Figure 7: Fixed πΆ1, πΆ2, πΆ3and π
1and simulated πΏ
1variation input
reflection coefficient.
to πΏ1. According to (9), the input matching circuit uses π
1
and πΆ3to reduce the number of inductors and make the π
factor smaller. π is inversely proportional to π 1. Therefore,
bandwidth is proportional to π 1. Figure 8 is fixed as πΆ
1
(9.51 pF), πΆ2(0.11 pF), πΆ
3(7.6 pF), and πΏ
1(0.91 nH), showing
that the bandwidth is proportional to the π 1. Figure 9 is fixed
as πΆ1(9.51 pF), πΆ
2(0.11 pF), πΆ
3(7.6 pF), and πΏ
1(0.91 nH),
showing that the noise figure is inversely proportional to theπ 1. ππis inversely proportional to πΆ
3while π is inversely
proportional to πΆ3. Therefore, the bandwidth is proportional
to the πΆ3. Figures 10 and 11 are fixed as πΆ
1(9.51 pF), πΆ
2
(0.11 pF), π 1(140Ξ©), and πΏ
1(0.91 nH), showing that the
bandwidth and noise figure are proportional to the πΆ3. The
size of the transistors π1, π 1, and πΆ
3must be carefully
selected. There is a trade-off in the design between widebandmatching and the noise figure. On the other hand, thedevice size must yield a sufficient noise performance andpower gain. As can be seen, the second-order filter ofπΏ1(0.91 nH), π
1(140Ξ©), and πΆ
3(7.6 pF) is employed as the
inputmatching network.The inputmatching network has lesscomplexity and better reflection coefficient from 3GHz to 10GHz.
3.2. Output Impedance Matching. Low-noise amplifiers relyon output impedance matching to achieve maximum powergain. A source follower technique has been widely used toprovide wideband output matching. Output impedance issimilar to 1/π
ππ , in which π
ππ is a gate-source transconduc-
tance of the source follower [4, 10]. However, it consumesmore power. For these reasons, we introduce an outputimpedance matching method suitable for wideband LNAdesign. Figure 12 shows the approximate output impedancesmall-signal equivalent circuit. The output impedance of theRLC network can be written as
Active and Passive Electronic Components 5
1 2 3 4 5 6 7 8 9 100 11Frequency (GHz)
R1 = 500 Ξ©R1 = 140 Ξ©R1 = 50 Ξ©
β20
β15
β10
β5
0
S 11
(dB)
Figure 8: Fixed πΆ1, πΆ2, πΆ3, and πΏ
1and simulated π
1variation input
reflection coefficient.
1 2 3 4 5 6 7 8 9 100 11
3456789
2
10
Frequency (GHz)
NF
(dB)
R1 = 500 Ξ©R1 = 140 Ξ©R1 = 50 Ξ©
Figure 9: Fixed πΆ1, πΆ2, πΆ3, and πΏ
1and simulated π
1variation noise
figure.
πout = ((ππ2//1
ππΆπ2
//ππΏ3) + ππΏ
4) //
1
ππΆ6
+1
ππΆ7
= π out + ππout,
(10)
πout
β(π+(1/π
π2πΆπ2)) ((πΆ
7+πΆ6) /πΆ6πΆ7)
π2+π (1/ππ2πΆπ2)+((πΆ
6πΏ4+πΆ6πΏ3+πΆπ2πΏ3) /πΆπ2πΆ6πΏ3πΏ4),
(11)
where πΆπ2
is the parasitic capacitance of π2at the drain
node, and ππ2
is the resistor of π2at the drain node. This
equivalent circuit can approximately be an RLC second-order
1 2 3 4 5 6 7 8 90 10 11Frequency (GHz)
β20
β15
β10
β5
0
C3 = 7.6 pFC3 = 0.95 pFC3 = 0.15 pF
S 11
(dB)
Figure 10: FixedπΆ1,πΆ2,π 1, and πΏ
1and simulatedπΆ
3variation input
reflection coefficient.
1 2 3 4 5 6 7 8 90
3456789
2
10
10 11
NF
(dB)
Frequency (GHz)C3 = 7.6 pFC3 = 0.95 pFC3 = 0.15 pF
Figure 11: FixedπΆ1,πΆ2,π 1, and πΏ
1and simulatedπΆ
3variation noise
figure.
filter structure. In (11), the quality factor of the filter circuitcan be given by
ππβ
1
2πβπΆ6πΏ4+ πΆ6πΏ3+ πΆπ2πΏ3
πΆπ2πΆ6πΏ3πΏ4
, (12)
π β ππ2πΆπ2βπΆ6πΏ4+ πΆ6πΏ3+ πΆπ2πΏ3
πΆπ2πΆ6πΏ3πΏ4
, (13)
where ππis the resonant frequency. In (12) and (13), in order
to obtain broader bandwidth, a low-quality factor needs tobe added to passive devices. π
πis inversely proportional to
πΆπ2
while bandwidth is inversely proportional to ππ2πΆπ2. The
πΆπ2capacitance is proportional to the width of the transistor.
The ππ2
resistance is inversely proportional to the width ofthe transistor. UWB system uses a specific frequency band
6 Active and Passive Electronic Components
Rs = 50 Ξ©
outZ
C7L4
C6L3rd2 Cd2
Figure 12: The wideband output matching small-signal equivalentcircuit.
2 3 4 5 6 7 8 9 10 11Frequency (GHz)
β30
β20
β25
β15
β10
β5
0
S 22
(dB)
M2 = 160/0.18 πmM2 = 240/0.18 πmM2 = 320/0.18 πm
Figure 13: Fixed πΆ6, πΆ7, πΏ3and πΏ
4, simulatedπ
2variation output
reflection coefficient.
(3.1β5GHz or 3.1β10.6GHz) to access data. Generally theresults of UWB return loss are less than β10 dB. The powerconsumption is proportional to the width of the transistor.According to (13), the narrowband LNA can be convertedinto a wideband amplifier by properly selecting π
2. This
provides good wideband matching and flat gain. Figure 13is fixed as πΆ
6(0.16 pF), πΆ
7(1.23 pF), πΏ
3(1.19 nH), and πΏ
4
(0.59 nH), showing that the simulated π2variation output
reflection coefficient. As can be seen, the second-order filterof π2(240/0.18 πm) is employed as the output matching
network. According to (12), the capacitor πΆ6and inductor
πΏ3-πΏ4will be optimized at the resonant frequency. Figure 14
is fixed as πΆ6(0.16 pF), πΆ
7(1.23 pF), π
2(240/0.18πm),
and πΏ4(0.59 nH), showing that the resonant frequency is
inversely proportional to the πΏ3. The output network has less
complexity and a better reflected coefficient from 3.1 GHz to10.6GHz.
3.3. Gain. At a high frequency, the current gain of commonsource configured transistor π
1is ππ1/π πΆgs1 [11]. Figure 15
shows the approximate first stage π1load of the simplified
2 3 4 5 6 7 8 9 10 11Frequency (GHz)
β20
β15
β10
β5
0
S 22
(dB)
L3 = 1.19nHL3 = 1.29nH
L3 = 1.09nH
Figure 14: Fixed πΆ6, πΆ7, π2, and πΏ
4and simulated πΏ
3variation
output reflection coefficient.
Id1
Vout1
C5
L2
Cd11/gm2
Figure 15: The first stage simplified small-signal equivalent circuit.
small-signal equivalent circuit [12]. The first stage load canbe approximated as
πload1 β1
ππΏ2πΆ5πΆπ1
β π2πΏ2πΆ5+ πππ2πΏ2+ 1
π2 + π (ππ2/πΆ5) + ((πΆ
5+ πΆπ1) / (πΏ2(πΆ5πΆπ1)))
.
(14)
The transfer function of the input matching network isπin(π ). The first stage voltage gain can be approximated as
π΄π1
β βππ1πin
ππΆgs1π π β πLoad1, (15)
where π π is the source resistance, in which voltage gain is
determined by the load inductor πΏ2, the total capacitance at
the drain of π1, and bypass capacitor πΆ
5. The second stage
voltage gain can be estimated as
π΄π2
β βππ2
ππΆgs2π in2β πout, (16)
Active and Passive Electronic Components 7
1 2 3 4 5 6 7 8 9 10
10
12
110
2
4
6
8
0
Frequency (GHz)
Ξ
K
Ξ K
Figure 16: πΎ (stability factor) and Ξ simulation result.
where π in2 is the input resistance at the gate ofπ2 and πoutis the output load impedance. The LNA voltage gain can beapproximated as
π΄πβππ1ππ2πinπLoad1πout
π2πΆgs1πΆgs2π π π in2. (17)
According to (17), the narrowband LNA can be convertedinto a wideband amplifier by properly selecting the devicesize. This provides good wideband matching, noise figureoptimization design, and flat gain.
3.4. Stability. Amplifier stability is important to preventnatural oscillation. The stability can be determined by the π-parameters, input and output matching network, and circuitterminations. The simpler tests show whether or not a devicecan be unconditionally stable [20]. One of these is the πΎ-Ξtest, which shows if a device can be unconditionally stable ifRolletβs condition is defined as
πΎ =1 β
π112
βπ22
2
+ |Ξ|2
2π12π21
> 1,
|Ξ| =π11π22 β π12π21
< 1.
(18)
The πΎ-Ξ test cannot be used to compare the relative sta-bility of twoormore devices because it involves constraints ontwo separate parameters. A new criterion has been proposedwhich combines the π-parameters in a test involving only asingle parameter, π, defined as
π =1 β
π112
π22 β Ξπβ
11
+π12π21
> 1. (19)
The performance of LNA is simulated using the advancedesign system (ADS). To consider the stability of the LNA,theπΎ factor andΞ should be considered. Figure 16 shows thatthe πΎ factor is always larger than 1 and the Ξ is smaller than1 all the time. Hence, this circuit is stable for all frequency
1 2 3 4 5 6 7 8 9 10 11
11
0Frequency (GHz)
3
5
7
9
1
πloadπsource
πlo
adπ
sour
ce
Figure 17: π factor simulation result.
RFin RFout
Vbias
DC power supply(HP 4142B)
Network analyzer(HP 8510C)
S-parameters test set(HP 8517B)
VDD
Figure 18: Test setup used for the LNA π-parameter measurements.
bands. There is another way to diagnose whether the LNAis unconditionally stable. The π factor at input and outputshould be larger than 1 in all frequency bands, as shown inFigure 17.
4. Measurement Results
On-wafer measurement is performed by an HP 8510C net-work analyzer and an HP 8517B is to test the π-parameter, asshown in Figure 18. A network analyzer is used to measurethe frequency response and input matching of the LNA.The input and output impedance matchings are both 50Ξ©.To ensure that the LNA still provides a conversion gainwhen process deviation occurs, the other process corners,namely, typical-NMOS typical-PMOS (TT), fast-NMOS fast-PMOS (FF), and slow-NMOS slow-PMOS (SS), are also usedto simulate this LNA. The results are shown in Figure 19.Figure 19 shows the measurement forward gain (π
21), from
3 to 5.6GHz; the measured gain is about 7β10 dB. Themeasurement data is 6 dB lower than the pre-simulation data
8 Active and Passive Electronic Components
201816141210
86420
2.8 3.3 3.8 4.3 4.8 5.3 5.8Frequency (GHz)
S 21
(dB)
Pre-simulation TTPost-simulation FFPost-simulation TT
Post-simulation SSMeasurement
Figure 19: Simulation and measurement results of conversion gainversus frequency of the proposed LNA. RF stands for simulationresults of modifiedMOSFET RFmodel. FF, TT, and SS represent thesimulation results of fast-NMOS fast-PMOS, typical-NMOS typical-PMOS, and slow-NMOS slow-PMOS process corners.
β10
0
β20
β30
β40
MeasurementSimulation
3 4 5 6 7 8 9 102 11Frequency (GHz)
S 11
(dB)
Figure 20: Measured and simulated input reflection coefficient.
because of the process drift and parasitic effect in the layout.The result of conversion gain measurement is situated in thepost-simulation (SS) process corner.
Figure 20 shows the input return loss (S11), from 3 to
11 GHz, and the measured S11
is less than β9 dB. Figure 21shows output return loss (S
22), from 3 to 7.5GHz, and the
measured S22is less thanβ8 dB. Figure 22 shows reverse isola-
tion (S12), from 3 to 11 GHz, and the measured S
12is less than
β40 dB. The simulation measured input of 1 dB compressionpoint at 5.3 GHz shown in Figure 23 is about β8 dB. Figure 24shows the measured fundamental output power and third-order intermodulation (IM3) for the RF input frequencyspacing of 1MHz, and the IIP
3is 2 dBm at 5.3 GHz. The
IM3 is measured, using two Agilnet E8247C continuous wave
β10
β15
β20
β25
β30
β5
0
MeasurementSimulation
3 4 5 6 7 8 9 102 11Frequency (GHz)
S 22
(dB)
Figure 21: Measured and simulated output reflection coefficient.
β10
β20
β30
β40
β50
β60
β70
β80
MeasurementSimulation
3 4 5 6 7 8 9 102 11
0
Frequency (GHz)
S 12
(dB)
Figure 22: Measured and simulated reverse isolation.
β40 β35 β30 β25 β20 β15 β10 β5 0 5 100
1
2
3
4
5
6
7
8
9
Gai
n (d
B)
Input power (dBm)
P1dB
Figure 23: Measured input 1 db compression point at 5.3 GHz.
Active and Passive Electronic Components 9
Table 2: Recently reported performance of UWB LNAs.
Reference Process CMOS(πm) π11 (dB)Avg. gain
(db) Freq. (GHz) NF (dB) IIP3 (dBm)Die area(mm2)
Power(mW) Topology
[4] 0.18πm CMOS
10 Active and Passive Electronic Components
reduces the chip area. References [10β12] only show core LNApower consumption, excluding the output source follower.References [14, 15] only show simulation. It should show thelinearity of the experiment. The LNA plays an important rolein improving overall system linearity. Table 2 clearly showsthat the proposed LNA has a very small chip area and thelowest power consumption.
5. Conclusion
An RLC wideband matching UWB LNA has been presentedin the above results, which can operate at a supply of 1 Vvoltage in a 0.18 πm CMOS technology. In the proposedtopology, the narrowband LNA can be converted into awideband amplifier by the RLC matching method. The RLCinput matching circuit reduces the chip area. The outputmatching method reduces power consumption. The mainadvantages of the LNA topology are low power use, moderatenoise, linearity, power gain, and a small chip area.
Conflict of Interests
The author indicated no potential conflict of interests.
Acknowledgments
The author would like to thank the National Science council(NSC) for funds support NSC99-2221-E-507-005 and theNational Chip Implementation Center (CIC) for technicalsupport.
References
[1] R. Harjani, J. Harvey, and R. Sainati, βAnalog/RF physical layerissues for UWB systems,β in Proceedings of the 17th InternationalConference on VLSI Design, pp. 941β948, January 2004.
[2] Z. Bai, S. Xu, W. Zhang, and K. Kwak, βPerformance analysis ofa novel m-ary code selected DS-BPAM UWB communicationsystem,β Journal of Circuits, Systems and Computers, vol. 15, no.3, pp. 455β466, 2006.
[3] Q. Yang,H. Zhu, andK. S. Kwak, βSuperimposed training-basedchannel estimation for MIMO-MB-OFDM UWB systems,βJournal of Circuits, Systems and Computers, vol. 17, no. 5, pp.865β870, 2008.
[4] J. Jung, T. Yun, and J. Choi, βUltra-wideband low noise amplifierusing a cascode feedback topology,β Microwave and OpticalTechnology Letters, vol. 48, no. 6, pp. 1102β1104, 2006.
[5] S. M. R. Hasan and J. Li, βA 12dB 0.7V 850W CMOS LNAfor 866MHz UHF RFID reader,β Active and Passive ElectronicComponents, vol. 2010, Article ID 702759, 5 pages, 2010.
[6] S. Toofan, A. R. Rahmati, A. Abrishamifar, and G. RoientanLahiji, βA low-power and high-gain fully integrated CMOSLNA,β Microelectronics Journal, vol. 38, no. 12, pp. 1150β1155,2007.
[7] B. M. Ballweber, R. Gupta, and D. J. Allstot, βA fully integrated0.5β5.5-GHz CMOS distributed amplifier,β IEEE Journal ofSolid-State Circuits, vol. 35, no. 2, pp. 231β239, 2000.
[8] X. Guan and C. Nguyen, βLow-power-consumption and high-gain CMOS distributed amplifiers using cascade of inductively
coupled common-source gain cells for UWB systems,β IEEETransactions on Microwave Theory and Techniques, vol. 54, no.8, pp. 3278β3283, 2006.
[9] C.-P. Chang and H.-R. Chuang, β0.18 πm 3-6 GHz CMOSbroadband LNA for UWB radio,β Electronics Letters, vol. 41, no.12, pp. 696β698, 2005.
[10] C.-W. Kim,M.-S. Kang, P. T. Anh, H.-T. Kim, and S.-G. Lee, βAnultra-wideband CMOS low noise amplifier for 3-5-GHz UWBsystem,β IEEE Journal of Solid-State Circuits, vol. 40, no. 2, pp.544β547, 2005.
[11] A. Bevilacqua and A. M. Niknejad, βAn ultrawideband CMOSlow-noise amplifier for 3.1-10.6-GHz wireless receivers,β IEEEJournal of Solid-State Circuits, vol. 39, no. 12, pp. 2259β2268,2004.
[12] M.-F. Chou, W.-S. Wuen, C.-C. Wu, K.-A. Wen, and C.-Y. Chang, βA CMOS low-noise amplifier for ultra widebandwireless applications,β IEICE Transactions on Fundamentals ofElectronics, Communications and Computer Sciences, vol. e88-A, no. 11, pp. 3110β3117, 2005.
[13] Y.-J. E. Chen and Y.-I. Huang, βDevelopment of integratedbroad-band CMOS low-noise amplifiers,β IEEE Transactions onCircuits and Systems I, vol. 54, no. 10, pp. 2120β2126, 2007.
[14] C. Wang, J. Jin, and F. Yu, βA CMOS 2-11 GHz continuousvariable gain UWB LNA,β IETE Journal of Research, vol. 56, no.6, pp. 367β372, 2010.
[15] A. Marzuki, A. Md. Shakaff, and Z. Sauli, βA 1.5 V, 0.85-13.35GHZ MMIC low noise amplifier design using optimizationtechnique,β IETE Journal of Research, vol. 55, no. 6, pp. 309β314,2009.
[16] A. Telli, S. Demir, and M. Askar, βCMOS planar spiral inductormodeling and low noise amplifier design,β MicroelectronicsJournal, vol. 37, no. 1, pp. 71β78, 2006.
[17] D. K. Shaeffer and T. H. Lee, βA 1.5-V, 1.5-GHz CMOS low noiseamplifier,β IEEE Journal of Solid-State Circuits, vol. 32, no. 5, pp.745β759, 1997.
[18] T. H. Lee, βThe design of CMOS radio-frequency integratedcircuits,β Cambridge, 2004.
[19] B. Razavi, Design of Analog CMOS Integrated Circuit, McGraw-Hill, 2001.
[20] D. M. Pozar,Microwave and RF Design of Wireless System, JohnWiley & Sons, 2001.
International Journal of
AerospaceEngineeringHindawi Publishing Corporationhttp://www.hindawi.com Volume 2014
RoboticsJournal of
Hindawi Publishing Corporationhttp://www.hindawi.com Volume 2014
Hindawi Publishing Corporationhttp://www.hindawi.com Volume 2014
Active and Passive Electronic Components
Control Scienceand Engineering
Journal of
Hindawi Publishing Corporationhttp://www.hindawi.com Volume 2014
International Journal of
RotatingMachinery
Hindawi Publishing Corporationhttp://www.hindawi.com Volume 2014
Hindawi Publishing Corporation http://www.hindawi.com
Journal ofEngineeringVolume 2014
Submit your manuscripts athttp://www.hindawi.com
VLSI Design
Hindawi Publishing Corporationhttp://www.hindawi.com Volume 2014
Hindawi Publishing Corporationhttp://www.hindawi.com Volume 2014
Shock and Vibration
Hindawi Publishing Corporationhttp://www.hindawi.com Volume 2014
Civil EngineeringAdvances in
Acoustics and VibrationAdvances in
Hindawi Publishing Corporationhttp://www.hindawi.com Volume 2014
Hindawi Publishing Corporationhttp://www.hindawi.com Volume 2014
Electrical and Computer Engineering
Journal of
Advances inOptoElectronics
Hindawi Publishing Corporation http://www.hindawi.com
Volume 2014
The Scientific World JournalHindawi Publishing Corporation http://www.hindawi.com Volume 2014
SensorsJournal of
Hindawi Publishing Corporationhttp://www.hindawi.com Volume 2014
Modelling & Simulation in EngineeringHindawi Publishing Corporation http://www.hindawi.com Volume 2014
Hindawi Publishing Corporationhttp://www.hindawi.com Volume 2014
Chemical EngineeringInternational Journal of Antennas and
Propagation
International Journal of
Hindawi Publishing Corporationhttp://www.hindawi.com Volume 2014
Hindawi Publishing Corporationhttp://www.hindawi.com Volume 2014
Navigation and Observation
International Journal of
Hindawi Publishing Corporationhttp://www.hindawi.com Volume 2014
DistributedSensor Networks
International Journal of